Circuits for intermediate-frequency-filterless, double-conversion receivers

ABSTRACT

Circuits for receivers including: N first mixers that each receive an input signal, are each clocked by a different phase of a first common clock frequency, and each provide an output; and for each N first mixer: a set of M second mixers, wherein each second mixer receives as an input the output of a same one of the N first mixers unique to the set, wherein each of M second mixer is clocked by a different phase of a second common clock frequency, and wherein each second mixer has an output; a set of M resistors having a first side and a second side, wherein the first side is connected to the output of a corresponding one of the set of M second mixers; and a set of M trans-impedance amplifiers that each having an input connected to the second side of a corresponding one of the resistors.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional Patent Application No. 63/130,070, filed Dec. 23, 2020, and is a continuation-in-part of U.S. patent application Ser. No. 17/176,349, filed Feb. 16, 2021, which claims the benefit of U.S. Provisional Patent Application No. 62/977,007, filed Feb. 14, 2020, and of U.S. Provisional Patent Application No. 63/130,070, filed Dec. 23, 2020, each of which is hereby incorporated by reference herein in its entirety.

BACKGROUND

The ever-increasing demands on wireless throughput require modern handset receivers to aggregate signals from multiple non-contiguously allocated RF carriers. Accordingly, new receivers that can receive signals from multiple non-contiguous RF carriers are desirable.

SUMMARY

In accordance with some embodiments, circuits for intermediate-frequency-filterless, double-conversion receivers are provided.

In some embodiments, circuits for a receiver are provided, the circuits comprising: N first mixers that each receive an input signal, that are each clocked by a different phase of a first common clock frequency, and that each provide an output, wherein N is a count of the first mixers; and for each of the N first mixers: a set of M second mixers, wherein M is a count of the second mixers in the set, wherein each second mixer in the set of M second mixers receives as an input the output of a same one of the N first mixers unique to the set, wherein each of the M second mixers in the set is clocked by a different phase of a second common clock frequency, and wherein each of the second mixers has an output; a set of M resistors having a first side and a second side, wherein the first side of each of the set of M resistors is connected to the output of a corresponding one of the set of M second mixers; and a set of M trans-impedance amplifiers that each having an input connected to the second side of a corresponding one of the set of M resistors and having an output.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is an example of a schematic of a receiver in accordance with some embodiments.

FIG. 2 is an example of a schematic of a mixer-first branch of a receiver in accordance with some embodiments.

FIG. 3 is an example of a schematic of second layer circuitry of a mixer-first branch of a receiver in accordance with some embodiments.

FIG. 4 is an example of a schematic of harmonic re-combination circuitry of a receiver in accordance with some embodiments.

FIG. 5 is an example of a schematic of sideband separation circuitry of a receiver in accordance with some embodiments.

FIG. 6 is an example of a schematic of a single-ended mixer first branch of a receiver in accordance with some embodiments.

FIG. 7 is an example of a schematic of a two dual-conversion low noise transconductance amplifier branches of a receiver in accordance with some embodiments.

FIG. 8 is an example of a schematic of a low noise transconductance amplifier of a receiver in accordance with some embodiments.

FIG. 9 is an example of Miller compensated trans-impedance amplifiers of a receiver in accordance with some embodiments.

FIG. 10 is an example of a table for sinusoidally modulating a transconductor of an in-phase branch of a receiver in accordance with some embodiments.

FIG. 11 is an example of a table for sinusoidally modulating a transconductor of a quadrature-phase branch of a receiver in accordance with some embodiments.

FIG. 12 is an example of a table for thermometer coding in accordance with some embodiments.

FIG. 13 is an example of a table for controlling transconductance unit cells in accordance with some embodiments.

FIG. 14 is an example of a block diagram of double-conversion, noise-cancelling receiver in accordance with some embodiments.

FIG. 15 is an example of a block diagram of a double-layer mixer-first branch in accordance with some embodiments.

FIG. 16A is an example of a timing diagram showing M non-overlapping clocks at a frequency F_(IF) in accordance with some embodiments.

FIG. 16B is an example of a timing diagram showing N non-overlapping clocks at a frequency F_(LO) in accordance with some embodiments.

FIG. 17 is an example of S11 graphs for fully singled-ended, single-ended-differential combination, and fully differential implementations of a double-layer mixer-first branch over different pairs of clock frequencies in accordance with some embodiments.

FIG. 18 is an example of model of a double-layer mixer-first branch in accordance with some embodiments.

FIG. 19 is an example of a block diagram of two quadrature-modulated low noise transconductance amplifier (LNTA) branches in accordance with some embodiments.

FIG. 20 is an example of a graph showing modulation of a transconductor in accordance with some embodiments.

FIG. 21 is an example of a behavioral model of modulated LNTAs in a single-ended-differential implementation in accordance with some embodiments.

FIG. 22 is an example of a model of part of an LNTA branch in accordance with some embodiments.

FIG. 23 is an example of a model of a modulated LNTA in accordance with some embodiments.

FIG. 24 is an example of graphs showing decomposition of a 16-phase modulated transconductance into four {−1,0,1} component waveforms in accordance with some embodiments.

FIG. 25 is an example of a model to study the effects of gain and phase imbalances on sideband rejection in the modulated LNTA branches in accordance with some embodiments.

FIG. 26 is an example of a simplified schematic for the (x, y)^(th) path of a double-conversion, noise cancelling receiver in accordance with some embodiments.

DETAILED DESCRIPTION

In accordance with some embodiments, circuits for intermediate-frequency-(IF)-filter-less, double-conversion receivers for concurrent dual-carrier reception are provided.

Turning to FIG. 1, in accordance with some embodiments, an example 100 of a schematic of a circuit for an intermediate-frequency-(IF)-filter-less, double-conversion receiver for concurrent dual-carrier reception is shown. As illustrated, circuit 100 includes a receiver front end 102 (which comprises two low-noise transconductance amplifier (LNTA) branches 104 and multiple double-conversion mixer-first branches 106), harmonic recombination and sideband separation circuitry 108, digital noise cancellation circuitry 110, a first clock source 114, and a second clock source 116.

In some embodiments, the circuit of FIG. 1 supports concurrent reception from two RF carriers have frequencies in a range from 100 MHz to 1200 MHz and separated apart by 200 MHz to 600 MHz.

In some embodiments, during operation in some modes, double-conversion mixer-first branches 106 translate a low-pass baseband impedance twice: first to a frequency F_(M); and then to a frequency (F_(C)+/−F_(M)). In some embodiments, doing this provides concurrent narrow-band impedance matching at two distinct frequencies only, while reflecting out-of-band signals for good linearity.

In some embodiments, LNTA branches 104 use direct digital synthesis (DDS)-modulated LNTAs for multi-phase, switched-transconductance mixing at F_(M), and standard 8-phase mixing at F_(C) with harmonic rejection (HR) baseband circuits.

In some embodiments, two RF carriers at (F_(C)+/−F_(M)) can be received, while spurious responses at (m·F_(C)+/−n·F_(M)) can be reduced for m<(M−1) (e.g., 7) and n<(N−1) (e.g., 15) with M-phase (e.g., 8-phase) F_(C) and N-phase (e.g., 16 phase) F_(M) clocks, where M and N are integers and powers of 2.

If some embodiments, this architecture can be extended to more clock phases to suppress more harmonics, subject to the process technology supporting the necessary clock speeds. For example, compared to 6 nm CMOS, a better process node (i.e., CMOS processes with smaller transistor feature lengths) (e.g., in 28 nm CMOS) usually offers a better logic gate for smaller gate delay and faster edge transition. Thus, in some embodiments, if one were to use 28 nn CMOS process, the DDS circuits can operate at a significantly higher clock speed to support more DDS clock phases.

As shown in FIG. 1, signals received at RF_(in) 112 can include real and imaginary components 122 and 124, respectively, at F_(C)-F_(M) and real and imaginary components 126 and 128, respectively, at (F_(C)+F_(M)). In response to these signals, circuit 100 can produce components 132, 134, 136, and 138 at the outputs of digital noise cancellation circuitry 110.

Although circuit 100 is shown in FIG. 1 as being implemented in a differential manner, it should be understood that circuit 100 can be implemented in a single-ended manner in some embodiments.

Turning to FIG. 2, an example 200 of a schematic of a circuit that can be used to implement four double-conversion mixer-first branches (MFBs) 106 and portions of circuitry 108 in accordance with some embodiments is shown. In some embodiments, circuit 200 creates an RF interface with tuned impedance matching at (F_(C)+/−F_(M)).

As illustrated, circuit 200 includes four first layer mixers 202, four second layer circuits 204, harmonic recombination circuits 206, sideband separation circuits 208, first clock source 210, second clock source 212, first 12.5% duty cycle clock generator 214, and second 12.5% duty cycle clock generator 216.

In some embodiments, RF signals around (F_(C)+/−F_(M)) are received at V_(RF), down-converted to F_(M), and then further down-converted to baseband without IF filtering. The eight baseband outputs from second layer circuits 204 are harmonically combined into four linearly independent outputs 242, 244, 246, and 248, while rejecting higher-order F_(C) harmonics. Addition and subtraction circuits then extract the I/Q components from each RF carrier to provide signals BB_(I-1), BB_(Q-1), BB_(Q-2), and BB_(I-2).

As described above in connection with FIG. 1, signals in V_(RF) received at mixers 202 can include real and imaginary components 122 and 124, respectively, at F_(C)-F_(M) and real and imaginary components 126 and 128, respectively, at F_(C)+F_(M). In response to these signals, circuitry 206 can produce components 222 and 226 (corresponding to components 122 and 126, respectively) at 242, components 224 and 228 (corresponding to components 124 and 128, respectively) at 244, components 224 and 228 (corresponding to components 124 and 128, respectively) at 246, and components 222 and 226 (corresponding to components 122 and 126, respectively) at 248. Components 232, 234, 238, and 236 can then provided at outputs BB_(I-1), BB_(Q-1), BB_(Q-2), and BB_(I-2), respectively.

In some embodiments, mixers 202 can be implemented in any suitable manner. For example, in some embodiments mixers 202 can be implemented using RF switches. In some embodiments, each RF switch can be realized as a custom-designed LVT RF NMOS transistor, placed in a deep N-well with the body terminal floating to ground.

In some embodiments, a switch width of 100 μm can be used for both the first-layer mixers (mixers 202) and the second-layer mixers (mixers 302 (see below)). In some embodiments, an alternate way to size the switches of mixers 202 and 302 is to use small-size switches for the F_(C) clock (mixers 202) and large-size switches for the F_(M) clock (mixers 302), such that the sum of the two switch resistances stays the same.

In some embodiments, each of first layer mixers 202 is clocked by a unique pair of phases (e.g.: phases 0 and 4; phases 1 and 5; phases 2 and 6; or phases 3 and 7) of an eight (0 . . . 7) phase, 12.5% duty cycle, non-overlapping clock at a frequency F_(C).

Although circuit 200 is shown in FIG. 2 as being implemented in a differential manner, it should be understood that circuit 200 can be implemented in a single-ended manner in some embodiments.

Turning to FIG. 3, an example 300 of a schematic of a circuit that can be used to implement each second layer circuit 204 of FIG. 2 in accordance with some embodiments is shown. As illustrated, circuit 300 includes four differential passive mixers 302 (each of which is connected to two of eight F_(M) clock phases as shown in the figure), a passive HR termination network 303, two differential transimpedance amplifiers (TIAs) 312, and four feedback capacitors CF.

Although circuit 300 is shown in FIG. 3 as being implemented in a differential manner, it should be understood that circuit 300 can be implemented in a single-ended manner in some embodiments.

In some embodiments, mixers 302 can be implemented in any suitable manner. For example, in some embodiments mixers 302 can be implemented using switches which can be custom-design LVT RF NMOS transistors, placed in a deep N-well with body terminals floating to ground.

In some embodiments, passive HR termination network 303 includes baseband capacitors C_(B) 304, and resistors 306, 307, 308, 309, 310, and 312. In some embodiments, resistors 306 and 308 can have values of 2*R_(B), resistors 307 and 309 can have values of (2+√{square root over (2)})*R_(B), resistors 310 can have values of 2*R_(B), and resistors 312 can have values of 2*√{square root over (2)}*R_(B), where R_(B) is any suitable value as described below. In some embodiments, C_(B) can have a value of 10 pF, CF can have a value of 3.5 pF (for single-carrier reception) or 0.89 pF (for dual-carrier reception), and RF can have a value of 4.5 kΩ (for single-carrier reception) or 18 kΩ (for dual-carrier reception).

In some embodiments, during operation, passive HR termination network 303 combines the down-converted signals with sinusoidal weighting in currents, while maintaining a constant resistance seen by the baseband capacitors C_(B). It rejects 3rd and 5th F_(M) harmonics at the input of baseband TIAs 312 and offers a tuned impedance matching at F_(M). By providing circuit 300 as the termination of each first-layer mixer 202, which uses a pair of an 8-phase differential passive mixers clocked at F_(C), the tuned RF interface is then translated to (F_(C)+/−F_(M)).

The narrow-bandpass tuned impedance matching at (F_(C)+/−F_(M)) reflects the out-of-band blocker signals, thus enhancing the out-of-band linearity of LNTA branches 104 significantly.

Turning to FIG. 4, components of harmonic recombination circuitry 206 are shown in accordance with some embodiments. As illustrated, circuitry 206 includes amplifiers 402 and 404, subtracters 406, and adders 408 in some embodiments. Any suitable amplifiers can be used to implement amplifiers 402 and 404, and amplifiers 402 and 404 can have gains of one and 1/√{square root over (2)}, respectively, in some embodiments. Any suitable subtracters and adders can be used to implement subtracters 406 and adders 408, respectively, in some embodiments.

Turning to FIG. 5, components of sideband separation circuitry 208 are shown in accordance with some embodiments. As illustrated, circuitry 208 includes adders 502 and subtracters 504 in some embodiments. Any suitable adders and subtracters can be used to implement adders 502 and subtracters 504, respectively, in some embodiments.

As described further below, in some embodiments, circuit 100 can be configured to operate in a variety of modes. For example, in some embodiments, circuit 100 can be configured for single-carrier reception or for concurrent, double-carrier reception.

In some embodiments, when the circuit of FIG. 1 is performing single-carrier reception, first-layer mixers 202 can be bypassed using any suitable circuitry (e.g., switches (not shown)), and second-layer mixers 302 can be clocked at F_(C) instead of at F_(M). In some such embodiments, four sets of second layer circuits 204 can be operated in parallel to help reduce the switch and routing resistance and improve the out-of-band S₁₁ reflection for better linearity, but at the cost of a dynamic power penalty. Alternatively, in some such embodiments, all but one second layer circuit can be turned off.

In some embodiments, when the circuit of FIG. 1 is performing concurrent, dual-carrier reception, the double-conversion mixer-first branches can be treated as two 8-path filters connected in series and terminated with low-pass, baseband impedances. These double-conversion mixer-first branches can be implemented in a fully single-ended, a single-ended-differential, or a fully differential realization.

In some embodiments, when the circuit of FIG. 1 is operating for single-carrier reception as described above and in a single-ended realization, its RF input impedance can be represented by:

$\begin{matrix} \begin{matrix} {{Z_{in}(\omega)} = {\frac{1}{4} \cdot \left\{ {{2R_{SW}} + {8 \cdot {\sum\limits_{m = {- \infty}}^{+ \infty}\;{{\alpha_{m}}^{2} \cdot {Z_{BB}\left( {\omega - {m \cdot \omega_{C}}} \right)}}}}} \right\}}} \\ {= {\frac{R_{SW}}{2} + {2 \cdot {\sum\limits_{m = {- \infty}}^{+ \infty}{{\alpha_{m}}^{2} \cdot {Z_{BB}\left( {\omega - {m \cdot \omega_{C}}} \right)}}}}}} \end{matrix} & (1) \end{matrix}$

where Z_(BB)(W) is the loading impedance, R_(SW) is the passive mixer switch resistance, m is any integer, |α_(m)|=|sinc(mπ/8)/8|, and we is 2πF_(C). For a source impedance of 50Ω and ideal mixer switches (i.e., R_(SW)=0), R_(B) needs to be 1.68 kΩ for impedance matching.

Similarly, in some embodiments, when the circuit of FIG. 1 is operating for single-carrier reception as described above and in a differential realization, its RF input impedance can be represented by:

$\begin{matrix} \begin{matrix} {{Z_{in}(\omega)} = {\frac{1}{4} \cdot \left\{ {{4R_{SW}} + {8 \cdot {\sum\limits_{m = {- \infty}}^{+ \infty}\;{{{2\alpha_{m}}}^{2} \cdot {Z_{BB}\left( {\omega - {m \cdot \omega_{C}}} \right)}}}}} \right\}}} \\ {= {R_{SW} + {2 \cdot {\sum\limits_{m = {- \infty}}^{+ \infty}{{{2\alpha_{m}}}^{2} \cdot {Z_{BB}\left( {\omega - {m \cdot \omega_{C}}} \right)}}}}}} \end{matrix} & (2) \end{matrix}$

where m is an odd integer. For a source impedance of 100Ω and ideal mixer switches, R_(B) needs to be 0.84 kΩ for impedance matching.

In some embodiments, when the circuit of FIG. 1 is operating for dual-carrier reception and in a fully single-ended double-conversion mixer-first branch, its RF input impedance can be represented as follows:

$\begin{matrix} {{Z_{in}(\omega)} = {{2R_{SW}} + {8^{2} \cdot {\sum\limits_{m = {- \infty}}^{+ \infty}\;{\sum\limits_{n = {- \infty}}^{+ \infty}\;{{\alpha_{m}}^{2} \cdot {\alpha_{n}}^{2} \cdot \cdot {Z_{BB}\left\lbrack {\omega - \left( {{m \cdot \omega_{C}} + {n \cdot \omega_{M}}} \right)} \right\rbrack}}}}}}} & (3) \end{matrix}$

where m, n are any integers, |α_(n)|=|sinc(nπ/8)/8|, we is 2πF_(C), and ω_(M) is 2 πF_(M). The input impedance is then twice the switch resistance in series with the scaled, frequency-translated baseband impedance at (m·F_(C)+n·F_(M)). For ideal mixer switches (i.e., R_(SW)=0), R_(B) needs to be 3.53 kΩ for impedance matching.

In some embodiments, the profiles have spurious matching at (m·F_(C)+n·F_(M)) where m and n are any integers. To reduce the spurious matching, the second-layer passive mixers can be realized differentially, given that the first-layer passive mixers produce differential outputs. The RF input impedance can thus be represented by:

$\begin{matrix} {{Z_{in}(\omega)} = {{2R_{SW}} + {\frac{8^{2}}{2} \cdot {\sum\limits_{m = {- \infty}}^{+ \infty}\;{\sum\limits_{n = {- \infty}}^{+ \infty}\;{{\alpha_{m}}^{2} \cdot {\alpha_{n}}^{2} \cdot \cdot \left\lbrack {1 + e^{{- j} \cdot {({m + n})} \cdot }} \right\rbrack^{2} \cdot {{Z_{BB}\left\lbrack {\omega - \left( {{m \cdot \omega_{C}} + {n \cdot \omega_{M}}} \right)} \right\rbrack}.}}}}}}} & (4) \end{matrix}$

where m and n are any integers. Impedance matching now occurs at (m·F_(C)+n·F_(M)), where (m+n) is even. For ideal mixer switches, R_(B) needs to be 1.76 kΩ for impedance matching.

With a differential realization of both the first-layer mixers and the second-layer mixers, the unwanted responses get suppressed for even m and n. The RF input impedance can thus be represented by:

$\begin{matrix} {{Z_{in}(\omega)} = {{4R_{SW}} + {\frac{8^{2}}{2} \cdot {\sum\limits_{m = {- \infty}}^{+ \infty}\;{\sum\limits_{n = {- \infty}}^{+ \infty}\;{{{2\alpha_{m}}}^{2}{{{2\alpha_{n}}}^{2} \cdot \cdot {Z_{BB}\left\lbrack {\omega - \left( {{m \cdot \omega_{C}} + {n \cdot \omega_{M}}} \right)} \right\rbrack}}}}}}}} & (5) \end{matrix}$

where m, n are both odd integers. For ideal mixer switches, R_(B) needs to be 0.88 kΩ for impedance matching, hence the R_(B) values for both single-carrier and concurrent dual-carrier reception are the same to the first order.

In some embodiments in which single-carrier reception is being performed as described above, the first-layer passive mixers are bypassed, and the four sets of second-layer mixers are operated in parallel and clocked at F_(C). In this case, the total switch resistance will be reduced by a factor of four. In some embodiments, FIG. 6 can be used to study gain and noise performance of this configuration. In some embodiments, the circuit of FIG. 6 performs the harmonic recombination at the baseband TIA inputs. The conversion gain can be derived as:

$\begin{matrix} {{{CG}_{{MFB},{SNGL}} \equiv \frac{V_{{MFB},I}}{V_{RF}}} = {\frac{G_{MXR} \cdot R_{F,{MFB}}}{R_{SW} + {\eta\; R_{B}}} = {\frac{1}{4} \cdot \frac{R_{F,{MFB}}}{R_{SW} + {\eta\; R_{B}}} \cdot {{\sin c}\left( {/8} \right)}}}} & (6) \end{matrix}$

where G_(MXR)=sinc(π/8)/4 is the passive mixer current conversion gain, R_(F,MFB) is the TIA feedback resistance, and η=8·|α₁|² is the impedance translation coefficient.

The noise factor of this configuration can be represented by:

$\begin{matrix} {F \approx {\frac{1}{{\sin c}^{2}\left( {/8} \right)} \cdot {\left\{ {1 + \frac{R_{SW}}{R_{S}} + {\frac{R_{B}}{8R_{S}} \cdot \left\lbrack {1 + {\sqrt{2} \cdot \left( \frac{R_{1} + R_{B}}{R_{B}} \right)^{2}}} \right\rbrack} + {\frac{\gamma}{G_{m,{op}}R_{S}} \cdot \left\lbrack {\frac{1}{4} + {\frac{R_{1} + R_{B}}{2} \cdot \left( {{\frac{\sqrt{2}}{2} \cdot \frac{1}{R_{B}}} + \frac{1}{R_{F,{MFB}}}} \right)}} \right\rbrack^{2}}} \right\}.}}} & (7) \end{matrix}$

For R_(S)=50Ω, R_(SW)=2.5Ω, R_(B)=399.2Ω, γ=1 (for 65 nm CMOS process), G_(m,op)=3 mS, R₁=(R_(S)+R_(SW)), and R_(F,MFB)=4.5 kΩ, the NF is calculated as 12.2 dB, whereas a simulated NF using schematic-level behavioral models can be calculated as 12.4 dB. The 0.2 dB difference probably stems from the power loss.

In some embodiments, a single-ended-differential realization yields the same performance as that of a fully differential realization. Thus, in some embodiments, the conversion gain from the RF input to the sideband-separated output for fully differential realization can be represented by:

$\begin{matrix} \begin{matrix} {{CG}_{{MFB},{DUAL}} = {{\frac{1}{{2R_{SW}} + {2{\eta^{2} \cdot R_{B}}}} \cdot G_{MXR}^{2}}{R_{F,{MFB}} \cdot 2 \cdot 2}}} \\ {= {\frac{1}{4} \cdot \frac{R_{F,{MFB}}}{{2R_{SW}} + {2{\eta^{2} \cdot R_{B}}}} \cdot {{\sin c}^{2}\left( {/8} \right)}}} \end{matrix} & (8) \end{matrix}$

where 2η² is the impedance translation coefficient of the single-ended-differential configuration in equation (4). The first factor of ‘2’ stems from the harmonically recombining gain for the F_(C) clocks, and the second factor of ‘2’ is the sideband separation gain. Under the impedance matching condition (i.e., 2R_(SW)+2η²·R_(B)=R_(S)), equation (8) reduces to:

$\begin{matrix} {{C\; G_{{MFB},{DUAL}}} = {\frac{1}{4} \cdot \frac{R_{F,{MFB}}}{R_{S}} \cdot {{\sin c}^{2}\left( {/8} \right)}}} & (9) \end{matrix}$

Following the same logic and procedures, the noise factor can be represented by:

$\begin{matrix} {F \approx {\frac{1}{{\sin c}^{4}\left( {/8} \right)} \cdot \left\{ {1 + \frac{2R_{SW}}{R_{S}} + {\frac{R_{B}}{\left( {8^{2}/2} \right) \cdot R_{S}} \cdot \left\lbrack {1 + {\sqrt{2} \cdot \left( \frac{R_{1} + R_{B}}{R_{B}} \right)^{2}}} \right\rbrack} + {\frac{\gamma}{G_{m,{op}}R_{S}} \cdot \frac{1}{4} \cdot \left\lbrack {\frac{1}{4} + {\frac{R_{1} + R_{B}}{2} \cdot \left( {{\frac{\sqrt{2}}{2} \cdot \frac{1}{R_{B}}} + \frac{1}{R_{F,{MFB}}}} \right)}} \right\rbrack^{2}}} \right\}}} & (10) \end{matrix}$

where R₁ now is (8²/2)·(R_(S)+2R_(SW)). For R_(S)=50Ω, 2R_(SW)=10Ω, R_(B)=1412Ω, γ=1, G_(m,op)=750 uS, and R_(F,MFB)=18 kΩ, the NF is calculated as 13.1 dB, whereas the simulated NF is 13.6 dB. The 0.5 dB difference probably stems from the power loss.

In some embodiments, due to the time-varying nature and the transparency of the passive mixers in the first and second layers, the mixer-first branches may exhibit harmonic folding and down-conversion. While undesired signals at clock harmonics are down-converted, a differential N-path realization as described herein in accordance with some embodiments can help to suppress the responses at the even clock harmonics. In some embodiments, the HR termination networks described herein combine the down-converted signals in current with sinusoidal weights to reject the responses at the odd clock harmonics, up to the 5th harmonic. Undesired signals at clock harmonics can fold to the desired signal band. The harmonic folding rejection ratio (HFRR), which is the ratio of the gain of the wanted RF signals to the gain of the unwanted RF signals that fold back to the desired signal band, can be represented by:

$\begin{matrix} {{H\; F\; R\; R_{({m,n})}} = {\frac{{\sin c}^{2}\left( {/8} \right)}{{{\sin c}\left( {{m}/8} \right)}{{\sin c}\left( {{n}/8} \right)}}}} & (11) \end{matrix}$

where m=8k₁±1, n=8k₂±1, and k₁, k₂∈

. In some embodiments, increasing the number of clock phases, especially for the F_(M) clocks, can be used to mitigate the harmonic folding, however, at the cost of reducing the maximum RF operating frequency and increasing the dynamic switch power. In some embodiments, 8 phases can be used for both F_(C) and F_(M) clocks for the double-conversion mixer-first branches.

Double-conversion LNTA branches are incorporated into the circuit of FIG. 1 to perform noise cancellation with the mixer-first branches for better receiver sensitivity.

Turning to FIG. 7, an example 700 of a schematic of circuitry that can be used to implement LNTA branches 104 and part of circuitry 108 of FIG. 1 in accordance with some embodiments is illustrated.

As shown, circuitry 700 includes LNTA branches 702 and 704, harmonic combination circuits 706 and sideband separation circuits 708.

As described above in connection with FIG. 1, signals in V_(RF) can include real and imaginary components 122 and 124, respectively, at F_(C)-F_(M) and real and imaginary components 126 and 128, respectively, at F_(C)+F_(M). In response to these signals, circuits 706 can produce components 742 and 746 (corresponding to components 122 and 126, respectively) at bb_(1(t)), components 744 and 748 (corresponding to components 124 and 128, respectively) at bb_(2(t)), components 744 and 748 (corresponding to components 124 and 128, respectively) at bb_(3(t)), and components 742 and 746 (corresponding to components 122 and 126, respectively) at bb_(4(t)). Components 752, 754, 758, and 756 can then be provided at outputs bb_(I-1), bb_(Q-1), bb_(Q-2), and bb_(I-2), respectively.

In some embodiments, to support concurrent signal reception, the LNTA branches combine conventional low-noise receiver design with direct digital synthesis (DDS) modulation.

In some embodiments, each LNTA can be include DDS circuits 710/720, 31 (or any other suitable number) transconductor unit slices 730/732, mixers 734/736, and filters 738/739. Any suitable transconductor unit slices can be used to implement slices 730/732 in some embodiments. In some embodiments, mixers 734/736 can be implemented similarly to mixers 202 of FIG. 2, as described above. Filters 738/739 can be implemented in any suitable manner, such as using TIAs with feedback capacitors similarly to what is illustrated in FIG. 3, in some embodiments.

Each DDS circuit 710/720 comprises a numerically controlled oscillator (NCO) 712/722, a phase accumulator 714/724, a 32-depth (or any other suitable size) memory 716/726, and a logic decoder 718/728.

Each NCO 712/722 can provide a clock output at a frequency (e.g., for 8-phase DDS modulation, the NCO can provide a clock frequency of 8*F_(M), and for 16-phase DDS modulation, the NCO can provide a clock frequency of 16*F_(M)) set by a hardware processor or any other suitable control mechanism (not shown).

Each phase accumulator 714/724 can accumulate a count based on the output of the corresponding NCO and a control input (not shown) that controls the rate (e.g., 1×, 2×, 4×, 8×, etc.) at which the accumulator increments its count (e.g., for 8-phase DDS modulation, the accumulator can have an increment of 4, and for 16-phase DDS modulation, the accumulator can have an increment of 2).

Each memory 716/726 can include a look-up table that contains data for sinusoidally modulating the transconductor unit slices. In some embodiments, this table can be created as shown in FIG. 10 (for in-phase) or FIG. 11 (for quadrature-phase). As shown, the tables can receive a 5-bit (or any other suitable size) input and provide a 5-bit (or any other suitable size) magnitude (“MAG”) output and a polarity (“POL”) bit.

Each logic decoder 718/728 can include a thermometer encoding table (e.g., such as the table of FIG. 12) for converting the 5-bit magnitude output by the corresponding memory 716/726 into a 31-bit (or any other suitable size) thermometer encoded output. Each logic decoder 718/728 can also include 31 (or any other suitable number) transconductor unit cell control tables (an example of which is shown in FIG. 13). Each bit of the thermometer encoded output can the drive its own transconductor unit cell control table. The bit of the thermometer encoded output can be used as an output enable (“oe”) input along with the corresponding polarity bit (“pol”) to drive the transconductor unit cell control table. As illustrated, in response to the input signals “oe” (output enable) and “pol” (polarity), each transconductor unit cell control table can provide five output signals ctl_t, ctl_sp_A, ctl_sn_A, ctl_sp_B, and ctl_sn_B, which can be used to control transmission gates 812 in FIG. 8 as shown of a corresponding transconductor unit cell.

In this way, during operation, transconductance unit cells 730/732 can be sinusoidally modulated at F_(M) by DDS circuits 710 and 720, in some embodiments.

Each LNTA branch operates as a multi-phase, switched-transconductance mixer to translate signals from (F_(C)±F_(M)) to F_(C) in some embodiments. In some embodiments, to reject 3rd and 5th F_(M) harmonics, the DDS phase accumulator increment can be set to 4 and the DDS circuits can be clocked at 8·F_(M). In some embodiments, to additionally reject 7th and 9th F_(M) harmonics, the DDS phase accumulator increment can be set to 2 and the DDS circuits can be clocked at 16·F_(M).

In some embodiments, the RF currents at the outputs of the transconductor unit cells (I_(RF,I) and I_(RF,Q) in FIG. 7) are translated from F_(C) to baseband using passive mixers driven by 8-phase 12.5%-duty-cycle, non-overlapping clocks at F_(C). The outputs of the four baseband TIAs are harmonically combined by circuits 706 to reject 3rd and 5th F_(C) harmonics.

The two LNTA branches, when modulated with in-phase and quadrature-phase sinusoidal DDSs 710 and 720, respectively, generate four outputs, bb₁(t), bb₂(t), bb₃(t), bb₄(t), at the output of harmonic recombination circuits 706 that contain overlapping but linearly independent I/Q components from the two RF carriers at (F_(C)±F_(M)). The I/Q components of each RF carrier are extracted using baseband addition and subtraction circuits 762, 764, 766, and 768 in sideband separation circuits 708. For example, by summing the outputs bb₁(t) and bb₄(t) with addition circuit 762, the in-phase component bb_(I-1) from the lower RF carrier is obtained.

The components of circuits 706 and 708 can be implemented in the same manner as corresponding components in FIGS. 4 and 5 as described above.

In some embodiments, for single-carrier reception, one LNTA branch can be disabled, and the DDS controls in the other LNTA branch can be fixed, so that the receiver operates as an 8-phase harmonic rejection (HR) receiver.

The conversion gain of each LNTA branch from RF input to baseband output when operating in a dual-carrier reception mode can be represented by:

$\begin{matrix} {{CG_{{LB},{DUAL}}} = {{\frac{1}{2} \cdot G_{m,{pk}}}{R_{F,{LB}} \cdot {{\sin c}\left( \frac{}{N} \right)} \cdot {{\sin c}\left( \frac{}{8} \right)}}}} & (12) \end{matrix}$

where N is the number of DDS modulation phases, G_(m,pk) is the peak LNTA transconductance, and R_(F,LB) is the TIA feedback resistance.

For single-carrier operation, the branch operates as an 8-phase HR receiver with a conversion gain given by:

$\begin{matrix} {{CG_{{LB},{{SNG}L}}} = {{\frac{1}{2} \cdot G_{m,{pk}}}{R_{F,{LB}} \cdot {{\sin c}\left( \frac{}{8} \right)}}}} & (13) \end{matrix}$

which is very close to equation (12) except for the sinc(π/N) multiplication factor. In some embodiments, the conversion gains for both modes are very close; for 8-phase modulation, the conversion gain in the dual-carrier reception mode is only 0.2 dB lower than the gain for single-carrier reception, while for 16-phase modulation, the conversion gain is only 0.1 dB lower.

The noise factor of the DDS-modulated LNTA branch with 8-phase modulation at F_(M) and 8-phase HR mixing at F_(C) can be represented by:

$\begin{matrix} {F_{LB} = {\frac{1}{{\sin c}^{4}\left( {/8} \right)} \cdot \left\{ {2 + {\frac{2\gamma}{G_{m,{pk}}R_{S}} \cdot \left\lbrack {1 + {2{\cos\left( \frac{}{4} \right)}}} \right\rbrack}} \right\}}} & (14) \end{matrix}$

where the first term of ‘2’ is due to the noise of R_(S) (the source resistance) and R_(T) (the termination resistance).

The noise factor with 16-phase DDS modulation at F_(M) and 8-phase HR mixing at F_(C) can be represented by:

$\begin{matrix} {F_{LB} = {\frac{1}{{{\sin c}^{2}\left( {/8} \right)} \cdot {{\sin c}^{2}\left( {/16} \right)}} \cdot {\left\{ {2 + {\frac{2\gamma}{G_{m,{pk}}R_{S}} \cdot \left\lbrack {\frac{1}{2} + {\cos\left( \frac{}{8} \right)} + {\cos\left( \frac{}{4} \right)} + {\cos\left( \frac{3}{8} \right)}} \right\rbrack}} \right\}.}}} & (15) \end{matrix}$

For the double-conversion LNTA branches, the harmonic rejection happens in both the F_(C) and F_(M) clock domains. To the first order, the harmonic rejection ratio (HRR) is obtained by multiplying two HRR expressions; e.g., when using 8-phase DDS modulation and 8-phase F_(C) clocks, the HRR at the sideband-separated outputs of the double-conversion LNTA branches at (m·F_(C)+n·F_(M)) is:

$\begin{matrix} {{HRR}_{({m,n})} = {\frac{\sin\;{c\left( {\pi/8} \right)}}{\sin\;{c\left( {m\;{\pi/8}} \right)}} \cdot \frac{\sin\;{c\left( {\pi/8} \right)}}{\sin\;{c\left( {n\;{\pi/8}} \right)}} \cdot \cdot \frac{1 + {{\rho_{c} \cdot 2}{\cos\left( {\pi/4} \right)}}}{1 + {{\rho_{c} \cdot 2}{\cos\left( {m\;{\pi/4}} \right)}}} \cdot \frac{1 + {{\rho_{m} \cdot 2}{\cos\left( {\pi/4} \right)}}}{1 + {{\rho_{m} \cdot 2}{\cos\left( {n\;{\pi/4}} \right)}}}}} & (16) \end{matrix}$

where m, n are both odd integers, ρ_(m) is the ratio of the quantized, mid-level transconductance and the peak transconductance, and ρ_(c) is the ratio of the baseband voltage gains used in the harmonic recombining network for the F_(C) clock.

In some embodiments, the mixer-first architecture with the incorporated, double-conversion LNTA branches as described herein can only cancel part of the noise of the baseband termination resistors shown in FIG. 3. Because of the configuration of the HR termination network, some of the noise appears with the same conversion polarity at the outputs of the two signal branches, and some appear in with an opposite conversion polarity. For example, the noise due to the shunt 2R_(B) and (2+√{square root over (2)})R_(B) resistors in the mixer-first branches will produce outputs with an opposite polarity, whereas the noise due to the series 2R_(B) and (2√{square root over (2)})R_(B) resistors produces outputs with the same polarity.

As described above, in some embodiments, the circuit of FIG. 1 can operate in a single-carrier reception mode. In this mode, the passive mixers of the LNTA branches are driven by the same 8-phase 12.5%-duty cycle, non-overlapping clocks at F_(C) as those for the mixer-first branches. The noise factor after cancellation when in this mode, F_(NC,SNGL), can be represented by:

$\begin{matrix} {F_{{NC},{SNGL}} = \frac{{\overset{\_}{v_{{no},{SNGL}}^{2}}/\Delta}\; f}{\begin{matrix} {{2 \cdot 4}{{kTR}_{S} \cdot \left\lbrack {\frac{R_{SW} + {\eta\; R_{B}}}{R_{S} + R_{SW} + {\eta\; R_{B}}} \cdot} \right.}} \\ \left. \left( {{CG}_{{LB},{SNGL}} - {K \cdot {CG}_{{MFB},{SNGL}}}} \right) \right\rbrack^{2} \end{matrix}}} & (17) \end{matrix}$

where v_(no,SNGL) ² /Δf is the total noise at the combined output, k is the Boltzmann constant, and T is temperature.

By properly selecting the value of K (the coefficient to adjust the relative gain of the two LNTA branches, which can be found by simulation), the noise due to any of the resistors is 303 of FIG. 3 and the baseband op-amps can be partially cancelled.

In some embodiments, when the circuit of FIG. 1 is operating in dual-carrier reception mode and is being modulated by 8-phase in-phase and quadrature-phase DDSs, the noise factor after sideband separation with noise cancellation, F_(NC,DUAL), can be represented by:

$\begin{matrix} {F_{{NC},{DUAL}} = \frac{{\overset{\_}{v_{{no},{DUAL}}^{2}}/\Delta}\; f}{\begin{matrix} {{2 \cdot 4}{{kTR}_{S} \cdot \left\lbrack {\frac{{2R_{SW}} + {2{\eta^{2} \cdot \; R_{B}}}}{R_{S} + {2R_{SW}} + {2{\eta^{2} \cdot \; R_{B}}}} \cdot} \right.}} \\ \left. \left( {{CG}_{{LB},{DUAL}} - {K \cdot {CG}_{{MFB},{DUAL}}}} \right) \right\rbrack^{2} \end{matrix}}} & (18) \end{matrix}$

where v_(no,DUAL) ² /Δf is the total noise at the combined output.

In some embodiments, the bandwidth at the RF input node should cover all significant higher-order harmonics (e.g, the 3rd, 5th, 7th, and 9th clock harmonics for 8-phase receivers) to avoid a large NF degradation. E.g., the bandwidth at the RF input node should be greater than 4900 MHz for F_(C)=700 MHz.

Turning to FIG. 8, an example 800 of a schematic of a differential modulated low noise transconductance amplifier (LNTA) that can be used to implement LNTAs 730 and 732 of FIG. 7 in accordance with some embodiments is shown.

As illustrated in FIG. 8, the differential modulated LNTA uses two cascoded common-source amplifiers 802 and 804. In some embodiments, there can be 31 (or any other suitable number) identical unit slices in each of the cascoded common-source amplifiers. As also shown, all 31 (or any other suitable number) of the unit slices share a central common-mode feedback circuit (comprising resistors 806, operational amplifier 808, and resistors 810) for stabilized DC operating points.

In some embodiments, the common-source devices can be sized for a (gm/ID) of 10 (or any other suitable number) for good linearity, and the cascoded devices can be sized for a (gm/ID) of 16 (or any other suitable number) for good noise performance.

In some embodiments, to enable or disable a slice rapidly during modulation, the output of each unit slice can be connected to a switch matrix (e.g., formed by transmission gates 812-822 in FIG. 8) that conducts the RF current to either the subsequent mixing stage (via transmission gates 814, 816, 820, and 822 controlled by ctl_sp_A<0:30>, ctl_sn_A<0:30>, ctl_sp_B<0:30>, ctl_sn_B<0:30>), respectively, or a dummy low-impedance termination (e.g., 0.6V at the output (o) of transmission gates 812 and 818 controlled by ctl_t<0:30>).

As shown in FIG. 8, each of transmission gates 812-822 can be formed as shown in 824.

In some embodiments, the operating frequency of each LNTAs is limited by the junction capacitances from drain and source terminals of the LNTA to the substrate. These capacitances stem from the cascoded devices and the switch matrices. In some embodiments, to mitigate these capacitances, the switches in all of the switch matrices after each unit slice can be designed with transmission gates using low-voltage CMOS technology (LVT) devices with floating bodies to rails. In some embodiments, for the same purpose, the 8-phase mixers can use transmission gates that are also floating their bodies to rails. In some embodiments, this approach can result in each mixer cell having a 20% reduction in parasitic capacitance with 8Ω switch resistance.

Turning to FIG. 9, an example 900 of a schematic of a Miller-compensated operational amplifier that can be used to implement baseband transimpedance amplifiers of low-noise transconductance amplifier (LNTA)branches 104 and multiple double-conversion mixer-first branches 106 of FIG. 1 in some embodiments is shown.

In some embodiments, the TIAs can use programmable feedback resistors and programmable feedback capacitors for gain control and bandwidth control, respectively.

In some embodiments, each TIA has an equivalent, differential 15 pF capacitor at its inputs to attenuate the down-converted, out-of-band blocking signals.

It is noted that in FIGS. 8 and 9, certain component sizes and voltages are shown. It should be understood that these sizes and voltages are merely for purposes of illustration and that any suitable component sizes and voltages can be used in some embodiments.

The trace routing resistance from the mixer outputs to the baseband TIA inputs limits the linearity of the signal branch. In some embodiments, multiple thin metal layers can be stacked to bring the routing resistance below 3Ω. This resistance can be further reduced with CMOS processes that offer more ultra-thick metal (UTM) layers in some embodiments.

In some embodiments, for the non-overlapping mixer clocks at F_(C), differential input clocks running at 4·F_(C) can be first divided by four using standard, 4-stage CMOS latches, producing 8-phase 50%-duty-cycle clocks, and then NOR logic gates can be used to generate the 8-phase 12.5%-duty-cycle, non-overlapping clocks. In some embodiments, the nonoverlapping mixer-clocks at F_(M) can be generated in the same way.

In some embodiments, to accommodate the need for different DDS clock rates, extra reconfigurable clock dividers can be used to support 8-phase and 16-phase DDS modulation with higher input clock rates.

In some embodiments, direct digital synthesizer circuits 710 and 720 in LNTA branches 702 and 704, respectively, are designed to vary the LNTA transconductances sinusoidally with a period of 1/F_(M). In some embodiments, direct digital synthesizer circuits 710 and 720 each contains a phase accumulator with programmable accumulating increments, a 7-bit-wide, 32-depth flip-flop-based SRAM as its look-up table, a thermometer-like logic decoder, and 31 drivers for each LNTA unit slice switch matrix.

In some embodiments, in the digital domain, gain and I/Q phase imbalances can be compensated and the signals then harmonically combined to reject 3rd and 5th F_(C) harmonics. Sideband separation can also be performed to extract I/Q information from each RF carrier in some embodiments.

In some embodiments, for concurrent dual-carrier reception, if, for example, the lower RF carrier is targeted, a single-point calibration can be performed by first injecting a continuous wave tone near the higher RF carrier with a 2 MHz intermediate frequency and acquiring the coefficients for gain and phase mismatches to cancel this tone at the low-band baseband output.

In some embodiments, more sophisticated compensation techniques, like multi-tap adaptive filtering, can be used for further improvement in harmonic rejection and sideband separation.

The resulting calibration coefficients can be used for measurements in some embodiments.

In some embodiments, noise cancellation can be realized by first performing complex baseband shifting and weighting to the mixer-first branch I/Q outputs and then summing these outputs with the LNTA branch outputs.

To cancel the termination noise from the mixer-first branches, standard mixer-first branches arranged in a double-conversion fashion can be used in some embodiments. More particularly, in some embodiments, the outputs of the second-layer, 8-phase mixer switches can be connected to corresponding input of a TIA (one for each mixer) each by a resistor R_(B), and harmonic recombination can be realized afterward. The noise due to these termination resistors at the outputs of the two signal branches may appear as common mode, whereas the desired signals may appear differential. Then, the termination noise can be fully cancelled, and the system's noise factor becomes:

$\begin{matrix} {F_{NC} \approx {\frac{1}{\sin\;{c^{4}\left( {\pi/8} \right)}} \cdot \left\{ {1 + {\frac{\gamma}{G_{m,{pk}}R_{S}} \cdot \ \left\lbrack {\frac{1}{2} + {\cos\left( \frac{\pi}{4} \right)}} \right\rbrack}} \right\}}} & (19) \end{matrix}$

In some embodiments, as the number of clock phases increases, the number of TIAs can also be increased. However, to maintain the same noise performance, the TIA operational amplifiers can be sized down, and the TIA feedback resistance can be sized up the same amount in some embodiments.

In some embodiments, more conversion stages can be used to receive more signals by putting one or more extra set of mixers before the first layer mixers. For example, to concurrently receive four carriers at (F_(C)±F_(M)±FN), three layers of passive mixing can be used in the mixer-first branch clocked at F_(C), F_(M), and FN with F_(C)>F_(M)>F_(N).

The low-pass, baseband impedance is then first converted to FN, then to (F_(M)±FN), and next to (F_(C)±F_(M)±FN), thus offering narrow-band tuned impedance matching at four distinct frequencies. Signals at those frequencies are down-converted to baseband and can be separated using addition and subtraction circuits. Similarly, more conversion stages can be included after the modulated LNTAs. In this case, the LNTAs are modulated at FN and are followed by two passive-mixing layers clocked at F_(M) and F_(C), respectively. However, more passive mixing layers require more series RF switches, resulting in a larger equivalent switch resistance and more complicated signal routing.

Turning to FIG. 14, an example simplified diagram of a double-conversion, noise-cancelling receiver 1400, featuring concurrent tuned matching, concurrent reception, noise cancellation, and rejection of spurious responses, in accordance with some embodiments is shown. As illustrated in FIG. 14, in some embodiments, first-layer passive mixers 1404 are connected to an RF input 1402 and clocked at F_(IF) by a clock 1406. In some embodiments, second-layer mixers 1408 are clocked at F_(IF) by a clock 1410 and loaded with low-pass, baseband impedances 1412. In some embodiments, this structure up-converts the baseband impedance first to F_(IF) and then to (F_(LO)±F_(IF)), resulting in tuned, high-Q bandpass RF impedance matching in two frequency bands and low input impedance elsewhere. Meanwhile, in some embodiments, the structure down-converts the RF carriers at these two frequencies and serves as a mixer-first receiver for concurrent signal reception, while rejecting harmonic responses without needing IF filters.

Turning to FIG. 15, a block diagram of a double-layer mixer-first branch 1500 that can be used in receiver 1400 and that uses two layers of passive mixing at F_(LO) and F_(IF) and provides narrow-band impedance matching at (F_(LO)±F_(IF)) while concurrently receiving I/Q information from these two RF carriers in accordance with some embodiments is shown.

The operation of double-layer mixer-first branch 1500 that creates an RF interface with tuned impedance matching at (F_(LO)±FT) simultaneously in accordance with some embodiments is now described. As shown in FIG. 15, double-layer mixer-first branch 1500 includes N second-layer mixing circuits 1502 in some embodiments. Each of N second-layer mixing circuits 1502 includes M-phase passive mixers 1504 clocked with non-overlapping, M-phase clocks ρ_(i)(t) 1506 (an example of which is illustrated in FIG. 16A) with frequency F_(IF), termination resistors R_(B) 1508, capacitors C_(B) 1510, M baseband TIAs 1512, followed by two F_(IF) harmonic recombining circuits 1514 and 1516, in some embodiments. IF harmonic recombining circuits 1514 and 1516 sum the weighted TIA output voltages to reject responses at higher-order F_(IF) harmonics, in some embodiments. Each set of second-layer mixers 1504 has a tuned input impedance centered at F_(IF), in some embodiments. In some embodiments, by using N sets of second-layer mixers (each set is in a circuit 1502) as the termination for the first-layer, N-phase passive mixers 1518 clocked with non-overlapping, N-phase clocks ξ_(j)(t) 1520 (an example of which is illustrated in FIG. 16B) with frequency F_(LO), the tuned RF interface is translated to (F_(LO)±F_(IF)) simultaneously as shown by graphs 1522. In some embodiments, N and M can be any suitable integer values. However, to simplify clock generation, in some embodiments, N and M can be integers that are powers of two (e.g., 4, 8, 16, 32, 64, etc.).

As described above in connection with FIG. 14, in some embodiments, the signal carriers around (F_(LO)±F_(IF)) in the RF input signal, V_(RF)(t), are first down-converted to F_(IF) at the output of mixers 1518 and then to baseband at the output of mixers 1504.

In some embodiments, the 2N baseband outputs the two IF harmonic recombination circuits 1514 and 1516 across each of the N circuits 1502 are then harmonically combined by LO harmonic recombination circuits 1522, 1523, 1524, and 1525 respectively into four baseband outputs, bb₀(t), bb₁(t), bb₂(t), and bb₃(t), while rejecting input signals around higher-order F_(LO) harmonics, in some embodiments. The I/Q components from each signal carrier, bb_(I/Q-1)(t) and bb_(I/Q-2)(t), at outputs 1530, 1531, 1532, and 1533 can be separated using addition and subtraction circuits 1526, 1527, 1528, and 1529 as shown in FIG. 15, in some embodiments.

In some embodiments, the double-layer mixer branch uses multi-phase F_(LO) clocks, such that the I/Q components of two RF carriers can be obtained from the linearly independent baseband outputs, bb₀(t), bb₁(t), bb₂(t), and bb₃(t), without any IF filtering, in some embodiments.

In some embodiments, double-layer mixer-first branch 1500 can be treated as an N-path filter, terminated with M-path filters that are loaded with low-pass, baseband impedances. In some embodiments, the first layer mixers 1518 and the second layer mixers 1504 can be implemented as both single-ended, as a single-ended-differential combination (i.e., mixers 1518 are single-ended and mixers 1504 are differential), or as both differential.

Z_(BB)(W) (FIG. 14) is the low-pass, baseband impedance, and z_(BB)(t) is its corresponding impulse response in some embodiments. It is assumed that TIAs 1512 provide a good virtual ground within the desired frequency band, such that Z_(BB)(W) is determined by R_(B) 1508 and C_(B) 1510, in some embodiments. Given the use of non-overlapping clocks, at any given moment, the RF current i_(RF)(t) flows into only one baseband path, in some embodiments. In some embodiments, the current for the (x, y)^(th) baseband path (where x is a value from 0 to N−1 and y is a value from 0 to M−1) is:

$\begin{matrix} {{i_{{BB}{({x,y})}}(t)} = {\left\lbrack {{\xi_{x}(t)}{\rho_{y}(t)}} \right\rbrack \cdot {{i_{RF}(t)}.}}} & (20) \end{matrix}$

This current then flows into z_(BB)(t) and produces the voltage:

$\begin{matrix} {{v_{{BB}{({x,y})}}(t)} = {\left\{ {\left\lbrack {{\xi_{x}(t)}{\rho_{y}(t)}} \right\rbrack \cdot {i_{RF}(t)}} \right\}*{z_{BB}(t)}}} & (21) \end{matrix}$

where * denotes convolution in some embodiments. In some embodiments, the voltage at the RF side of mixers 1518, V_(RF)(t), at any given moment, is equal to the voltage across the appropriate (x, y)^(th) baseband impedance plus the ohmic drop across two mixer switches in series:

$\begin{matrix} {{v_{RF}(t)} = {{2{R_{SW} \cdot {i_{RF}(t)}}} + {\sum\limits_{x = 0}^{N - 1}{\sum\limits_{y = 0}^{M - 1}{\left\lbrack {{\xi_{x}(t)}{\rho_{y}(t)}} \right\rbrack \cdot \left\langle {\left\{ {\left\lbrack {{\xi_{x}(t)}{\rho_{y}(t)}} \right\rbrack \cdot {i_{RF}(t)}} \right\}*{z_{BB}(t)}} \right\rangle}}}}} & (22) \end{matrix}$

where R_(SW) is the switch resistance, which is assumed to be equal for both layers of mixers 1518 and 1504. In some embodiments, the Fourier series of ξ_(x)(t) is:

$\begin{matrix} {{\xi_{x}(t)} = {\sum\limits_{k = {- \infty}}^{+ \infty}{\alpha_{k}{\exp\left( {{- {jxk}}\frac{2\pi}{N}} \right)}{\exp\left( {{jk}\;\omega_{LO}t} \right)}}}} & (23) \end{matrix}$

where α_(k)=(1/N)sinc(kπ/N) exp(−jkπ/N), k is any integer, and the Fourier series of ρ_(y)(t) is:

$\begin{matrix} {{\rho_{y}(t)} = {\sum\limits_{l = {- \infty}}^{+ \infty}{\beta_{l}{\exp\left( {{- {jyl}}\frac{2\pi}{N}} \right)}{\exp\left( {{jl}\;\omega_{IF}t} \right)}}}} & (24) \end{matrix}$

where β₁=(1/M)sinc(π/M) exp(−jπ/M). In some embodiments, using properties of the Fourier series, the summation term in (22) is:

$\begin{matrix} {{\mathcal{F}\left\langle {\left\lbrack {{\xi_{x}(t)}{\rho_{y}(t)}} \right\rbrack \cdot \left\{ {{\left\lbrack {{\xi_{x}(t)}{\rho_{y}(t)}} \right\rbrack \cdot {i_{RF}(t)}}*{z_{BB}(t)}} \right\}} \right\rangle} = {\sum\limits_{k = {- \infty}}^{+ \infty}{\sum\limits_{l = {- \infty}}^{+ \infty}{\sum\limits_{p = {- \infty}}^{+ \infty}{\sum\limits_{q = {- \infty}}^{+ \infty}{\alpha_{k}\beta_{l}\alpha_{p}{\beta_{q} \cdot {\exp\left\lbrack {{- {{jk}\left( {k + p} \right)}}\frac{2\pi}{N}} \right\rbrack}}{{\exp\left\lbrack {{- {{jy}\left( {l + q} \right)}}\frac{2\pi}{M}} \right\rbrack} \cdot {I_{RF}\left\lbrack {\omega - {\left( {k + p} \right)\omega_{LO}} - {\left( {l + q} \right)\omega_{IF}}} \right\rbrack} \cdot {Z_{BB}\left\lbrack {\omega - \left( {{p\;\omega_{LO}} + {q\;\omega_{IF}}} \right)} \right\rbrack}}}}}}}} & (25) \end{matrix}$

Now, in some embodiments, the Fourier transform of V_(RF)(t) is obtained as:

$\begin{matrix} {{V_{RF}(\omega)} = {{2{R_{SW} \cdot {I_{RF}(\omega)}}} + {{NM} \cdot {\sum\limits_{k = {- \infty}}^{+ \infty}{\sum\limits_{l = {- \infty}}^{+ \infty}{\sum\limits_{p = {- \infty}}^{+ \infty}{\sum\limits_{q = {- \infty}}^{+ \infty}{\alpha_{k}\beta_{l}\alpha_{p}{\beta_{q} \cdot {I_{RF}\left\lbrack {\omega - {\left( {k + p} \right)\omega_{LO}} - {\left( {l + q} \right)\omega_{IF}}} \right\rbrack} \cdot {Z_{BB}\left\lbrack {\omega - \left( {{p\;\omega_{LO}} + {q\;\omega_{IF}}} \right)} \right\rbrack}}}}}}}}}} & (26) \end{matrix}$

where (k+p)=k₁N, (1+q)=k₂M, and k₁, k₂∈

. In some embodiments, the input impedance Z_(in)(ω) can be found by ignoring other frequency components except for (k+p)=0 and (l+q)=0. V_(RF)(ω) becomes a function of only I_(RF)(ω) and:

$\begin{matrix} {{{Z_{in}(\omega)} \equiv \frac{V_{RF}(\omega)}{I_{RF}(\omega)}} = {{2R_{SW}} + {{NM} \cdot {\sum\limits_{p = {- \infty}}^{+ \infty}{\sum\limits_{q = {- \infty}}^{+ \infty}{{\alpha_{p}}^{2}{{\beta_{q}}^{2} \cdot {Z_{BB}\left\lbrack {\omega - \left( {{p\;\omega_{LO}} + {q\;\omega_{IF}}} \right)} \right\rbrack}}}}}}}} & (27) \end{matrix}$

FIG. 17 shows a comparison of analytical and simulated S₁₁ profiles for different mixer-first branch implementations for F_(LO)=700 MHz and R_(SW)=1.5Ω but different F_(IF) clock rates when N=M=8, in some embodiments. In some embodiments, for a fully single-ended implementation, R_(S)=50Ω, R_(B)=3.34 kΩ, and C_(B)=5 pF; for the single-ended-differential implementation, R_(S)=50Ω, R_(B)=1.67 kΩ, and C_(B)=10 pF; and for the fully differential implementation, R_(S)=100Ω, R_(B)=0.83 kΩ and C_(B)=20 pF.

As can be seen in the top row of FIG. 17, in some embodiments, the S₁₁ profile for the fully single-ended implementation has the desired impedance matching at (F_(LO)±F_(IF)) but also has spurious matching at (pF_(LO)+qF_(IF)) where p, q∈

.

In some embodiments, to achieve better Si′ profiles with less spurious matching, the first-layer passive mixers while single-ended can be configured to produce differential outputs and the second-layer passive mixers can be realized in a differential manner. This results in a single-ended-differential combination implementation. In such an implementation, in some embodiments, the RF input impedance would be:

$\begin{matrix} {{Z_{in}^{\prime}(\omega)} = {{2R_{SW}} + {\frac{NM}{2} \cdot {\sum\limits_{p = {- \infty}}^{+ \infty}{\sum\limits_{q = {- \infty}}^{+ \infty}{{\alpha_{p}}^{2}{{\beta_{q}}^{2} \cdot \left\{ {1 + {\exp\left\lbrack {{- {j\left( {p + q} \right)}}\pi} \right\rbrack}} \right\}^{2} \cdot {Z_{BB}\left\lbrack {\omega - \left( {{p\;\omega_{LO}} + {q\;\omega_{IF}}} \right)} \right\rbrack}}}}}}}} & (28) \end{matrix}$

To distinguish different expressions for different implementations (e.g., for input impedance, gain, and noise), (·)′ is used for single-ended-differential implementations and (·)″ is used for fully differential implementations, whereas the expressions without these symbols are for the fully single-ended implementations.

As can be seen in the middle row of FIG. 17, in some embodiments, the number of frequencies where spurious matching occurs reduces significantly for a single-ended-differential combination implementation. However, undesired impedance matching still happens at (pF_(LO)+qF_(IF)) where (p+q) is even, in some embodiments.

In some embodiments, a differential implementation for both the first- and the second-layer mixers further improves the Si′ profiles; the unwanted matching gets suppressed for even p and q. In some embodiments, the differential RF input impedance is:

$\begin{matrix} {{Z_{in}^{''}(\omega)} = {{4R_{SW}} + {\frac{NM}{2} \cdot {\sum\limits_{p = {- \infty}}^{+ \infty}{\sum\limits_{q = {- \infty}}^{+ \infty}{{{2\alpha_{p}}}^{2}{{{2\beta_{q}}}^{2} \cdot {Z_{BB}\left\lbrack {\omega - \left( {{p\;\omega_{LO}} + {q\;\omega_{IF}}} \right)} \right\rbrack}}}}}}}} & (29) \end{matrix}$

where p, q are both odd integers. As shown in the bottom row of FIG. 17, in some embodiments, impedance matching now occurs for (F_(LO)±FT) as desired with a few sets of undesired responses (e.g., (F_(LO)±3F_(IF))) creating significant matching within the practical bandwidth, whereas a low input impedance exists for all other frequencies.

Therefore, in some embodiments, for receiver systems allowing the use of RF input baluns, which therefore can provide a differential signal to the first layer mixers, it may be desirable to use a fully-differential implementation for its good matching profiles. For example, in some embodiments, handset receivers using Global System for Mobiles (GSM) and Code Division Multiple Access (CDMA) technologies can use differential RF inputs to make use of common-mode rejection and to leverage the shrinking voltage headroom, and therefore can be implemented using a fully-differential implementation as described herein.

In some embodiments, for receiver systems that do not allow the use of baluns (e.g., due to the limited form factor or system complexity), and which therefore may not provide a differential signal to the first layer mixers, it may be desirable to use a single-ended-differential implementation. For example, in some embodiments, handset receivers using Long-Term Evolution (LTE) and New Radio (NR) technologies can use single-ended RF inputs due to a higher number of supported bands (especially for Carrier Aggregation (CA)), a limited number of package pins, and the cost for differential matching networks, and therefore can be implemented using a single-ended-differential combination implementation as described herein.

In some embodiments, there is a trade-off between “large” and “small” mixer-switch sizing. In a “large”-switch design, the mixers may have a smaller R_(SW) and a larger parasitic switch capacitance than in a “small” switch design. Accordingly, for a given bandwidth, the resistors R_(B) can be sized larger, and the capacitors C_(B) can be smaller, in some embodiments. Smaller R_(SW) results in a lower out-of-band impedance and thus better out-of-band signal reflection, and such up-front filtering profile protects the LNTA branches from strong out-of-band blocking signals, in some embodiments. However, this choice faces design challenges, such as larger parasitic switch capacitance and higher switch-clock dynamic power that may be present in some embodiments. Using chip manufacturing processes with reduced parasitics (e.g., silicon on insulator (SOI)) can significantly mitigate those challenges in some embodiments. In a “small”-switch mixer design (i.e., with larger R_(SW)), these challenges are mitigated to some extent, in some embodiments. However, in some embodiments, to maintain the impedance matching with the same bandwidth, R_(B) needs to be smaller (due to the larger R_(SW)), which requires larger C_(B).

As discussed previously, for a single-ended RF input, the single-ended-differential combination implementation has a better matching profile compared to its fully single-ended counterpart. Therefore, below, details of the single-ended-differential combination implementation is discussed in conjunction with FIG. 18, in some embodiments. In some embodiments, the conversion gain of the single-ended-differential combination implementation is:

$\begin{matrix} \begin{matrix} {{{CG}_{MFB}^{\prime} \equiv \frac{V_{{BB},{1 - 1}}}{V_{RF}}} = {{\frac{R_{F,{MFB}}}{{2R_{SW}} + {2\eta\; R_{B}}} \cdot G_{{MX},{LO}}}{G_{{MX},{IF}} \cdot \frac{NM}{2}}}} \\ {= {{\frac{1}{2} \cdot \frac{R_{F,{MFB}}}{{2R_{SW}} + {2\eta\; R_{B}}} \cdot \sin}\;{{c\left( {\pi/N} \right)} \cdot \sin}\;{c\left( {\pi/M} \right)}}} \end{matrix} & (30) \end{matrix}$

where R_(F,MFB) is the TIA feedback resistance; G_(MX,LO)=(1/N)sinc(π/N) and G_(MX,IF)=(1/M)sinc(π/M) are the current conversion gains of the passive mixers driven by F_(LO) and F_(IF) clocks, respectively; η is the impedance translation coefficient and can be derived from (27) as:

$\begin{matrix} {\eta = {\frac{\sin\;{c^{2}\left( {\pi/N} \right)}}{N} \cdot \frac{\sin\;{c^{2}\left( {\pi/M} \right)}}{M}}} & (31) \end{matrix}$

R_(B) can be left as a design parameter, such that (30) is a generalized gain expression, in some embodiments. In some embodiments, if impedance matching to the antenna source resistance, R_(S), is desired, R_(B)′ can be selected using the following equation in some embodiments:

$\begin{matrix} {R_{B}^{\prime} = \frac{R_{S} - {2R_{SW}}}{2\eta}} & (32) \end{matrix}$

For example, in some embodiments, for N=M=8, R_(S)=50Ω, and R_(SW)=10Ω, R_(B)′ can be selected as 1.06 kΩ for input matching.

Similarly, in some embodiments, the conversion gain of a differential implementation can be expressed as:

$\begin{matrix} {{CG}_{MFB}^{''} = {{\frac{R_{F,{MFB}}}{{4R_{SW}} + {8\eta\; R_{B}}} \cdot \sin}\;{{c\left( {\pi/N} \right)} \cdot \sin}\;{c\left( {\pi/M} \right)}}} & (33) \end{matrix}$

where the impedance translation coefficient is 8η.

For matching to R_(S), R_(B)″ can be found using that following equation, in some embodiments:

$\begin{matrix} {R_{B}^{''} = \frac{R_{S} - {2R_{SW}}}{8\eta}} & (34) \end{matrix}$

For example, in some embodiments, when N=M=8, R_(SW)=10Ω, and R_(S)=100Ω, R_(B)″ can be selected as 0.53 kΩ for input matching.

From (30) and (33), as N, M increases, both sinc factors approach to unity, leading to better noise and harmonic performance, in some embodiments.

For well-designed receivers, in some embodiments, R_(S), R_(SW), R_(B), and the base-band op-amps in the TIAs may be the dominant sources of noise, while R_(F,MFB) should not contribute significant noise. In some embodiments, the baseband TIA typically offers a good virtual ground at the baseband frequencies of interest, which simplifies the analysis (since the TIA's input impedance can be ignored).

Regarding the noise from R_(S) and R_(SW): In some embodiments, these noise sources have a transfer function to the branch output similar to that of the desired signals, except that noise folding needs to be accounted for. In some embodiments, the output noise contribution due to R_(S) is:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},R_{S}}^{2}}}{\Delta\; f} = {\frac{\overset{\_}{v_{n,R_{S}}^{2}}}{\Delta\; f} \cdot \left( {\frac{{2R_{SW}} + {2\eta\; R_{B}}}{R_{S} + {2R_{SW}} + {2\eta\; R_{B}}} \cdot {CG}_{{MFB},{N - M}}^{\prime}} \right)^{2} \cdot \frac{2}{\sin\;{{c^{2}\left( {\pi/M} \right)} \cdot \sin}\;{c^{2}\left( {\pi/N} \right)}}}} & (35) \end{matrix}$

where v_(n,R) _(S) ² R_(S)/Δf=4 kT R_(s), k is the Boltzmann constant, and the following factor:

$\left( {\frac{{2R_{SW}} + {2\eta\; R_{B}}}{R_{S} + {2R_{SW}} + {2\eta\; R_{B}}} \cdot {CG}_{{MFB},{N - M}}^{\prime}} \right)^{2}$

accounts for the voltage division between R_(S) and the in-band input resistance R_(in)=2R_(SW)+2ηR_(B); in the following factor:

$\frac{2}{\sin\;{{c^{2}\left( {\pi/M} \right)} \cdot \sin}\;{c^{2}\left( {\pi/N} \right)}}$

the numerator of ‘2’ accounts for the noise down-conversion from both lower and upper sidebands around the RF carrier, whereas the sinc factors in the denominator model the noise folding from higher-order intermodulation products of the F_(LO) and F_(IF) clocks.

Regarding the noise from R_(B): In some embodiments, each path has a termination resistor, R_(B). Given the non-overlapping nature of the F_(LO) and F_(IF) clocks, in some embodiments, the noise from one signal path does not propagate to the other paths, so they are orthogonal in time, in some embodiments. Since the resistors are physically different, their noise is uncorrelated in some embodiments. Therefore, the noise from one path can be studied, and then the noise powers can be summed-up for all paths with corresponding weights for harmonic recombination and sideband separation.

FIG. 18 shows an example simplified schematic of the (x, y)^(th) path, in some embodiments. The noise of R_(B) can be modelled with a series voltage source, which only conducts noise current when ξ_(x)(t)·ρ_(y)(t) is high or when ξ_(x+N/2)(t)·ρ_(y+M/2)(t) is high, in some embodiments. The average resistance looking back into the mixer network at baseband, R₁, is then:

$\begin{matrix} {R_{1} = {\frac{NM}{2} \cdot \left( {R_{S} + {2R_{SW}}} \right)}} & (36) \end{matrix}$

Since ξ_(x)(t) and ρ_(y)(t) have duty cycles of 1/N and 1/M, respectively, ξ_(x)(t)·ρ_(y)(t) has a duty cycle of 1/(NM) for the period whose value is the inverse of the least common multiple (LCM) of the F_(LO) and the F_(IF) clock frequencies. Similarly, ξ_(x+N/2) (t)·ρ_(y+M/2)(t) also has a duty cycle of 1/(NM) over the same period. Within the desired, baseband channel frequencies, C_(B) is open and R₁ becomes (NM/2)·(R_(S)+2R_(SW)). At higher frequencies, C_(B) can be considered as a short circuit to ground, in some embodiments.

Thus, the output noise due to R_(B) in the (x, y)^(th) path is:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},{R_{B}{({x,y})}}}^{2}}}{\Delta\; f} = {\frac{\overset{\_}{v_{n,R_{B}}^{2}}}{\Delta\; f} \cdot \left\lbrack {\frac{R_{F,{MFB}}}{R_{B} + R_{1}} \cdot {\cos\left( {{\frac{2\pi}{N}x} - {\frac{2\pi}{M}y}} \right)}} \right\rbrack^{2}}} & (37) \end{matrix}$

where v_(n,R) _(B) ² /Δf=4 kT R_(B), and the cos factor is the coefficient due to harmonic recombination and sideband separation, in some embodiments.

Utilizing the orthogonal and uncorrelated properties, in some embodiments, the total output noise due to all R_(B)'S is then:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},R_{B}}^{2}}}{\Delta\; f} = {{\sum\limits_{x = 0}^{{N/2} - 1}{\sum\limits_{y = 0}^{M - 1}\frac{\overset{\_}{v_{{no},{R_{B}{({x,y})}}}^{2}}}{\Delta\; f}}} = {\frac{\overset{\_}{v_{n,R_{B}}^{2}}}{\Delta\; f} \cdot \frac{NM}{4} \cdot \left( \frac{R_{F,{MFB}}}{R_{B} + R_{1}} \right)^{2}}}} & (38) \end{matrix}$

Regarding the noise from baseband op-amps in the TIAs: In some embodiments, the noise of the baseband op-amps in the TIAs can be modelled as noise voltage sources at their non-inverting input (FIG. 18), and the analysis is similar to that for R_(B). In some embodiments, the output noise due to the op-amp in the (x, y)^(th) path is:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},{{op}{({x,y})}}}^{2}}}{\Delta\; f} = {\frac{\overset{\_}{v_{n,{op}}^{2}}}{\Delta\; f} \cdot \left\lbrack {\left( {1 + \frac{R_{F,{MFB}}}{R_{B} + R_{1}}} \right) \cdot {\cos\left( {{\frac{2\pi}{N}x} - {\frac{2\pi}{M}y}} \right)}} \right\rbrack^{2}}} & (39) \end{matrix}$

where v_(n,op) ² /Δf=4 kTγ/G_(m,op). In some embodiments, the total output noise due to all the baseband op-amps is then:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},{op}}^{2}}}{\Delta\; f} = {{\sum\limits_{x = 0}^{{N/2} - 1}{\sum\limits_{y = 0}^{M - 1}\frac{\overset{\_}{v_{{no},{{op}{({x,y})}}}^{2}}}{\Delta\; f}}} = {\frac{\overset{\_}{v_{n,{op}}^{2}}}{\Delta\; f} \cdot \frac{NM}{4} \cdot \left( {1 + \frac{R_{F,{MFB}}}{R_{B} + R_{1}}} \right)^{2}}}} & (40) \end{matrix}$

The noise factor of the mixer-first branch in a single-ended-differential realization, F_(MFB,N-M)′, can be derived, in some embodiments, by comparing the total output noise with the output noise due to R_(S) as follows:

$\begin{matrix} {F_{{MFB},{N - M}}^{\prime} = {\frac{\frac{\overset{\_}{v_{{no},R_{S}}^{2}}}{\Delta\; f} + \frac{\overset{\_}{v_{{no},R_{SW}}^{2}}}{\Delta\; f} + \frac{\overset{\_}{v_{{no},R_{B}}^{2}}}{\Delta\; f} + \frac{\overset{\_}{v_{{no},{op}}^{2}}}{\Delta\; f}}{2 \cdot \frac{\overset{\_}{v_{n,R_{S}}^{2}}}{\Delta\; f} \cdot \left( {\frac{{2R_{SW}} + {2\eta\; R_{B}}}{R_{S} + {2R_{SW}} + {2\eta\; R_{B}}} \cdot {CG}_{{MFB},{N - M}}^{\prime}} \right)^{2}} \approx {\frac{1}{\sin\;{{c^{2}\left( {\pi/N} \right)} \cdot \sin}\;{c^{2}\left( {\pi/M} \right)}} \cdot \left\{ {1 + \frac{2R_{SW}}{R_{S}} + {\frac{R_{B}}{R_{S}} \cdot \frac{2}{NM}} + {\frac{\gamma}{G_{m,{op}}R_{S}} \cdot \frac{2}{NM} \cdot \left\lbrack {1 + \frac{{\left( {R_{S} + {2R_{SW}}} \right) \cdot \left( {{NM}/2} \right)} + R_{B}}{R_{F,{MFB}}}} \right\rbrack^{2}}} \right\}}}} & (41) \end{matrix}$

The double-sideband noise factor is used here since its value is the same as the single-sideband noise factor after image rejection is performed, in some embodiments.

In some embodiments, the image rejection can be performed in the digital domain using the down-converted I/Q baseband signals.

In some embodiments, the sinc factors approach unity when the numbers of clock phases increase, and when N=M=8, the noise factor becomes:

$\begin{matrix} {F_{{MFB},{8 - 8}}^{\prime} \approx {\frac{1}{\sin\;{c^{4}\left( {\pi/8} \right)}} \cdot \left\{ {1 + \frac{2R_{SW}}{R_{S}} + {\frac{R_{B}}{32 \cdot R_{S}} \cdot \frac{\gamma}{{32 \cdot G_{m,{op}}}R_{S}} \cdot \left\lbrack {1 + \frac{{32 \cdot \left( {R_{S} + {2R_{SW}}} \right)} + R_{B}}{R_{F,{MFB}}}} \right\rbrack^{2}}} \right\}}} & (42) \end{matrix}$

Similarly, in some embodiments, the noise factor of a fully-differential realization, F_(MFB,N-M)″, for N=M=8 can be derived as:

$\begin{matrix} {F_{{MFB},{8 - 8}}^{''} \approx {\frac{1}{\sin\;{c^{4}\left( {\pi/8} \right)}} \cdot \left\{ {1 + \frac{4R_{SW}}{R_{S}} + {\frac{R_{B}}{8 \cdot R_{S}} \cdot \frac{\gamma}{{8 \cdot G_{m,{op}}}R_{S}} \cdot \left\lbrack {1 + \frac{{8 \cdot \left( {R_{S} + {4R_{SW}}} \right)} + R_{B}}{R_{F,{MFB}}}} \right\rbrack^{2}}} \right\}}} & (43) \end{matrix}$

Because of the time-varying nature of the passive mixers, mixer-first designs in some embodiments may face challenges of harmonic folding, meaning that undesired signals at clock harmonics can fold to the desired signal band. For standard, single-layer mixer-first branches using N-phase clocks at F_(LO), the harmonic folding rejection ratio (HFRR) is the ratio of the gain of the wanted RF signals to the gain of the unwanted RF signals that fold back on top of the desired signal band:

$\begin{matrix} {{HFRR}_{n} = {\frac{\sin\;{c\left( {\pi/N} \right)}}{\sin\;{c\left( {n\;{\pi/N}} \right)}}}} & (44) \end{matrix}$

where n=kN±1 and k∈

, and k is any integer. For the double-layer mixer-first branch, in some embodiments, to the first order, its HFRR can be obtained by multiplying two HFRR expressions:

$\begin{matrix} {{HFRR}_{n,m} = {{\frac{\sin\;{c\left( {\pi/N} \right)}}{\sin\;{c\left( {n\;{\pi/N}} \right)}} \cdot \frac{\sin\;{c\left( {\pi/M} \right)}}{\sin\;{c\left( {m\;{\pi/M}} \right)}}}}} & (45) \end{matrix}$

where n=k₁N±1, m=k₂M±1, and k₁, k₂∈

. Increasing the number of clock phases, especially for the Fw clocks, can be used to mitigate harmonic folding in some embodiments. For example, in some embodiments, when N=M=8, F_(LO)=700 MHz and F_(IF)=150 MHz, the response at |F_(LO)−9F_(IF)|=650 MHz will be folded back to the lower-carrier baseband output, whereas the response at |F_(LO)−7F_(IF)|=350 MHz will be folded back to the higher-carrier baseband output. Using a larger M (e.g., 16 and higher) eliminates these two responses but at the cost of reducing the maximum RF operating frequency and increasing the dynamic switch power due to the parasitics from the switching devices, in some embodiments.

In some embodiments, from equation (41), while the number of baseband branches increases as N, M increases, the overall noise performance can be kept constant by scaling down the individual TIA op-amps and their feedback capacitors, and scaling up their feedback resistance. This is because noise adds in power, whereas signal adds in voltage.

When the number of clock phases increases, designers can choose to keep the mixer-switch sizes the same or choose to reduce the mixer-switch size in some embodiments. In the latter case, the mixer-clock dynamic power may stay constant to the first order since the total switch size remains the same, in some embodiments. However, smaller mixer switches may have larger R_(SW), resulting in a higher out-of-band impedance and less out-of-band blocker filtering, in some embodiments. To maintain good out-of-band filtering, in some embodiments, the switches may be kept at the same size; however, as their number increases with the number of clock phases, the mixer-clock dynamic power may increase and require stronger clock buffers, in some embodiments.

Frequency translations in high-performance, current-mode receivers are usually realized by converting the RF voltage to current with LNTAs and then translating the RF information to baseband with passive mixers in the current domain. In accordance with some embodiments, if the LNTA transconductance is periodically modulated, another frequency translation can be realized during the RF voltage-to-current conversion.

In FIG. 19, in accordance with some embodiments, two quadrature-modulated LNTA branches 1902 and 1904 are shown, where each branch includes an M-phase-modulated LNTA 1906 and N-phase, current-mode mixers 1908, followed by baseband TIAs 910 and two harmonic recombination circuits 1912 and 1914. In some embodiments, N and M can be any suitable positive integer values. More particularly, in some embodiments, N and M can be integers that are powers of two (e.g., 4, 8, 16, 32, 64, etc.).

As shown in FIG. 20, in accordance with some embodiments, the two modulated LNTAs can have sinusoidally-varying transconductances, G_(m,I)(t) and G_(m,Q)(t), in quadrature at F_(IF). The two modulated LNTAs can operate as switched-G_(m) mixers to translate V_(RF)(ω) at (F_(LO)±F_(IF)) to I_(RF,I)(ω) and I_(RF,Q)(ω) at F_(LO), which are then translated to baseband with passive mixers driven at F_(LO). These baseband currents can be converted to voltages with TIAs and are further harmonically re-combined to form four baseband signals, V_(BB0)(ω) to V_(BB3)(ω), while rejecting higher-order F_(LO) harmonics, in some embodiments. The I/Q components from both RF carriers can be simply separated from these four baseband signals using addition and subtraction circuits, in some embodiments.

FIG. 21 is an example of a behavioral model of modulated LNTAs in a single-ended-differential implementation in accordance with some embodiments.

In some embodiments, the transconductance conversion gain of the modulated LNTAs, G_(m,EQ), can be defined as the ratio of I_(RF,I)(ω) at F_(LO) to V_(RF)(ω) at (F_(LO)±F_(IF)) and is the fundamental Fourier series coefficient of G_(m,I)(t):

$\begin{matrix} {{G_{m,{EQ}} \equiv \frac{I_{{RF},I}}{V_{RF}}} = {{\frac{1}{2} \cdot G_{m,{pk}} \cdot \sin}\;{c\left( {\pi/M} \right)}}} & (46) \end{matrix}$

where M is the number of the LNTA modulation phases, and G_(m,pk) is the peak LNTA transconductance. To derive closed-form expressions for gain and noise performance, it can be assumed that both G_(m,I)(t) and G_(m,Q)(t) are the discrete-time approximations of the sinusoids with un-quantized transconductance, in some embodiments. I_(RF,I)(ω) at F_(LO) is then translated and converted to the voltage V_(BB0)(ω) at baseband by the transimpedance conversion gain is:

$\begin{matrix} \begin{matrix} {R_{EQ} \equiv \frac{V_{{BB}\; 0}}{I_{{RF},I}}} \\ {= {G_{{MX},{LO}}{R_{F,{LB}} \cdot \frac{N}{2}}}} \\ {= {{\frac{1}{2} \cdot R_{F,{LB}} \cdot \sin}\;{c\left( {\pi/N} \right)}}} \end{matrix} & (47) \end{matrix}$

where R_(F,LB) is the TIA feedback resistance, and G_(MX,LO)=(1/N)sinc(π/N) is the mixer current conversion gain, in some embodiments. Here un-quantized, baseband weightings in the harmonic rejection circuits are also assumed. After sideband separation, in some embodiments, the conversion gain doubles and is:

$\begin{matrix} \begin{matrix} {{CG}_{{LB},{N - M}}^{\prime} \equiv \frac{V_{{BB},{I - 1}}}{V_{RF}}} \\ {= {G_{m,{EQ}}{R_{EQ} \cdot 2}}} \\ {= {{\frac{1}{2} \cdot G_{m,{pk}}}{R_{F,{LB}} \cdot \sin}\;{{c\left( {\pi/N} \right)} \cdot \sin}\;{{c\left( {\pi/M} \right)}.}}} \end{matrix} & (48) \end{matrix}$

As N, M increase, both sinc factors approach unity, and there will be less noise folding from higher-order harmonics and better harmonic suppression across the RF spectrum, in some embodiments. In some embodiments, if needed, a fully-differential implementation can be used to suppress common-mode interferences; its conversion gain is the same as that for the single-ended-differential implementation.

In some embodiments, input impedance matching in the receiver is provided by the double-layer mixer-first branches. For a noise analysis of the LNTA branches of FIG. 19, in some embodiments, the impedance matching of the double-layer mixer first branches can be modeled with a resistor R_(T) (as shown in FIG. 19) that is equal to R_(S) for broadband matching. In a well-designed current-mode receiver, the LNTAs can have a high output impedance in some embodiments.

In some embodiments, R_(S), R_(T), and the modulated LNTAs are the significant noise sources, while the noise from passive mixers, TIA feedback resistors, and TIA op-amps do not significantly contribute to the overall noise.

Regarding the noise from R_(S) and R_(T): These noise sources share the same noise transfer function to the branch output, in some embodiments. In some embodiments, the noise contribution for R_(S) is:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},{RS}}^{2}}}{\Delta\; f} = {\frac{\overset{\_}{v_{n,{RS}}^{2}}}{\Delta\; f} \cdot \left( \frac{{CG}_{{LB},{N - M}}^{\prime}}{2} \right)^{2} \cdot \frac{2}{\sin\;{{c^{2}\left( {\pi/N} \right)} \cdot \sin}\;{c^{2}\left( {\pi/M} \right)}}}} & (49) \end{matrix}$

where v_(nR) _(S) ² /Δf=4 kT R_(S), the factor of (1/2) stems from the fact that the noise process experiences a voltage division between R_(S) and R_(T) at the LNTA's input, and the two sinc factors are derived from noise folding.

Regarding the noise from modulated LNTAs: To understand the noise of modulated LNTAs in the overall signal branch, consider circuit 2200 in FIG. 22, in which modulated LNTA 2202 is directly followed by a baseband TIA 2204. Circuit 2200 down-converts the signals at F_(IF) to baseband, while rejecting other higher-order F_(IF) clock harmonics.

As shown in FIG. 23, in some embodiments, modulated LNTA 2202 can be implemented in some embodiments as a set of switched unit cells 2302. The noise of each unit cell 2302 can be modelled as a shunt noise current source 2304 at the cell output as shown in FIG. 23, in some embodiments. If a unit cell is used during LNTA modulation, its noise current appears at the TIA virtual ground and develops a noise voltage at the TIA output, in some embodiments. Otherwise, if the cell is not used during modulation, it does not contribute noise, in some embodiments.

Since the noise of each unit cell is un-correlated, these unit cells can then be re-organized, in some embodiments. Their equivalent, noise power spectral densities (PSDs) can be calculated from the sum of the individual noise PSDs, in some embodiments. Thanks to the sinusoidal symmetry, one can then: decompose G_(m,I)(t) into four components, f₁(t) to f₄(t), as shown in FIG. 24, with corresponding sinusoidal weights; compute the noise contribution of each component; and sum these contributions, in some embodiments. Using Parseval's theorem, the noise PSD can be derived from the integral of the square of a time-domain function over its period, in some embodiments. In some embodiments, the noise contribution at TIA output due to f₁(t) is:

$\begin{matrix} \begin{matrix} {{\frac{\overset{\_}{v_{n,f_{1}}^{2}}}{\Delta\; f} = {4{kT}\;\gamma\; G_{m,{pk}}{{\cos\left( \frac{3\pi}{8} \right)} \cdot \frac{1}{T_{IF}}}{\int_{0}^{T_{IF}}{{{f_{1}(t)}}^{2}{{dt} \cdot R_{F,{LB}}^{2}}}}}}\ } \\ {= {4{kT}\;\gamma\; G_{m,{pk}}{{\cos\left( \frac{3\pi}{8} \right)} \cdot \frac{7}{8} \cdot {R_{F,{LB}}^{2}.}}}} \end{matrix} & (50) \end{matrix}$

Similarly, in some embodiments, the contributions of the other components can be computed; all four contributions can be summed to find the total noise PSD:

$\begin{matrix} {\frac{\overset{\_}{v_{n,G_{m}}^{2}}}{\Delta\; f} = {{\sum\limits_{k = 1}^{4}\frac{\overset{\_}{v_{n,f_{k}}^{2}}}{\Delta\; f}} = {4k\; T\;\gamma\;{G_{m,{pk}} \cdot R_{F,{LB}}^{2} \cdot {{\frac{1}{8}\left\lbrack {1 + {2\;{\cos\left( \frac{\pi}{8} \right)}} + {2\;{\cos\left( \frac{\pi}{4} \right)}} + {2\;\cos\;\left( \frac{3\pi}{8} \right)}} \right\rbrack}.}}}}} & (51) \end{matrix}$

This expression includes the noise converted not only from the fundamental clock frequency at F_(IF), but also from higher-order clock harmonics (e.g., 15th and 17th F_(IF) harmonics), in some embodiments. In some embodiments, this 16-phase result can be generalized to an M-phase modulated LNTA, where in practice M=2^(k) with k E Z and k≥2, by decomposing its transconductance waveform into M/4 components and summing their contributions:

$\begin{matrix} {\frac{\overset{\_}{v_{n,G_{m}}^{2}}}{\Delta\; f} = {4\; k\; T\;\gamma\; G_{m,{pk}}{R_{F,{LB}}^{2} \cdot \frac{2}{M}}{\sum\limits_{k = 0}^{{M/2} - 1}{{{\cos\left( {\frac{2\pi}{M}k} \right)}}.}}}} & (52) \end{matrix}$

For convenience, equation (52) can be referred to the TIA input as:

$\begin{matrix} {\frac{\overset{\_}{i_{n,G_{m}}^{2}}}{\Delta\; f} = {4\; k\; T\;\gamma\;{G_{m,{pk}} \cdot \frac{2}{M}}{\sum\limits_{k = 0}^{{M/2} - 1}{{{\cos\left( {\frac{2\pi}{M}k} \right)}}.}}}} & (53) \end{matrix}$

Getting back to the overall signal branch in FIG. 19, the quadrature-modulated LNTAs 1906 are followed by N-phase passive mixers 1908, baseband TIAs 1910, and harmonic-recombining circuits 1912 and 1914. In some embodiments, extra noise may be down-converted from higher-order F_(LO) harmonics. In terms of noise process, the signal branch down-converts the noise from (nF_(LO)+mF_(IF)) to baseband, where n=k₁N±1, m=k₂M±1, and k1, k2∈

. In some embodiments, the resulting total noise at V_(BB,I-1) in FIG. 19 is then:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},G_{m}}^{2}}}{\Delta\; f} = {2 \cdot \frac{\overset{\_}{i_{n,G_{m}}^{2}}}{\Delta\; f} \cdot R_{EQ}^{2} \cdot \frac{2}{\sin\;{c^{2}\left( {\pi/N} \right)}}}} & (54) \end{matrix}$

where the factor of ‘2’ is due to the fact that two modulated LNTAs are used, and the sinc factor stems from the noise folding from the higher-order F_(LO) clock harmonics. Substituting (53) and (47) into (54), in some embodiments, the following is obtained:

$\begin{matrix} {{\frac{\overset{\_}{v_{{no},G_{m}}^{2}}}{\Delta\; f} = {{\frac{\overset{\_}{v_{n,G_{m,{pk}}}^{2}}}{\Delta\; f} \cdot \left( {CG}_{{LB},{N - M}}^{\prime} \right)^{2} \cdot \frac{4}{M}}{\sum\limits_{k = 0}^{{M/2} - 1}{{{\cos\left( {\frac{2\pi}{M}k} \right)}} \cdot \frac{2}{\sin\;{{c^{2}\left( {\pi/M} \right)} \cdot \sin}\;{c^{2}\left( {\pi/M} \right)}}}}}}{{{where}\mspace{14mu}{v_{n,G_{m,{pk}}}^{2}/\Delta}\; f} = {4{kT}\;{\gamma/{G_{m,{pk}}.}}}}} & (55) \end{matrix}$

In some embodiments, the noise factor of the quadrature-modulated LNTA branches, F_(LB,N-M)′, can now be derived as:

$\begin{matrix} {F_{{LB},{N - M}}^{\prime} = {\frac{\frac{\overset{\_}{v_{{no},R_{S}}^{2}}}{\Delta\; f} + \frac{\overset{\_}{v_{{no},R_{T}}^{2}}}{\Delta\; f} + \frac{\overset{\_}{v_{{no},G_{m}}^{2}}}{\Delta\; f}}{2 \cdot \frac{\overset{\_}{v_{{no},R_{S}}^{2}}}{\Delta\; f} \cdot \left( {\frac{1}{2}{CG}_{{LB},{N - M}}^{\prime}} \right)^{2}} = {\frac{1}{\sin\;{{c^{2}\left( {\pi/M} \right)} \cdot \sin}\;{c^{2}\left( {\pi/M} \right)}} \cdot \left\lbrack {2 + {{\frac{4\gamma}{G_{m,{pk}}R_{S}} \cdot \frac{4}{M}}{\sum\limits_{k = 0}^{{M/2} - 1}{{\cos\left( {\frac{2\pi}{M} \cdot k} \right)}}}}} \right\rbrack}}} & (56) \end{matrix}$

If 8-phase-modulated LNTAs are used with 8-phase HR down-conversion circuits (i.e., N=M=8), in some embodiments, (56) reduces to:

$\begin{matrix} {F_{{LB},{8 - 8}}^{\prime} = {\frac{1}{\sin\;{c^{4}\left( {\pi/8} \right)}} \cdot \left\{ {2 + {\frac{2\gamma}{G_{m,{pk}}R_{S}} \cdot \left\lbrack {1 + {2\;{\cos\left( \frac{\pi}{4} \right)}}} \right\rbrack}} \right\}}} & (57) \end{matrix}$

Similarly, in some embodiments, the noise factor of a fully-differential implementation, F_(LB,N-M)″, for N=M=8 can be derived as:

$\begin{matrix} {F_{{LB},{8 - 8}}^{''} = {\frac{1}{\sin\;{c^{4}\left( {\pi/8} \right)}} \cdot \left\{ {2 + {\frac{4\gamma}{G_{m,{pk}}R_{S}} \cdot \left\lbrack {1 + {2\;{\cos\left( \frac{\pi}{4} \right)}}} \right\rbrack}} \right\}}} & (58) \end{matrix}$

In some embodiments, the previous parts assume that the modulated LNTAs produce un-quantized transconductance, and that the harmonic recombination networks apply un-quantized weights to bb_(I,i)(t) and bb_(Q,i)(t). In practice, both weights are realized in a quantized manner. For modulated LNTAs, quantization errors result in undesired harmonic responses at higher-order F_(IF) harmonics. The harmonic rejection ratio (HRR) is the ratio of transconductance conversion gain at F_(IF) to the transconductance conversion gain at the i^(th) harmonic:

$\begin{matrix} {{{HRR}_{i} \equiv {\frac{G_{m,{EQ}}}{G_{m,i}}}} = {{\frac{\frac{1}{T_{IF}}{\int_{0}^{T_{IF}}{{G_{m,I}(t)}{\exp\left( {{1 \cdot j}\;\omega_{IF}t} \right)}{dt}}}}{\frac{1}{T_{IF}}{\int_{0}^{T_{IF}}{{G_{m,I}(t)}{\exp\left( {{i \cdot j}\;\omega_{IF}t} \right)}{dt}}}}}.}} & (59) \end{matrix}$

For M=8, in some embodiments, (59) reduces to

$\begin{matrix} {{HRR}_{i} = {{\frac{{sinc}\left( {\pi/8} \right)}{{sinc}\left( {i\;{\pi/8}} \right)}} \cdot {\frac{1 + {2{\rho_{IF} \cdot {\cos\left( {\pi/4} \right)}}}}{1 + {2{\rho_{IF} \cdot {\cos\left( {i\;{\pi/4}} \right)}}}}}}} & (60) \end{matrix}$

where ρ_(IF) is the ratio of the mid-level transconductance to the peak LNTA transconductance and should be cos(π/4), ideally. Due to quantization errors, it will deviate from this ideal value, resulting in a finite HRR and undesired signals around the F_(IF) clock harmonics being down-converted on top of the desired signals, in some embodiments. For 4-bit resolution, the modulated LNTAs offer 36.7 dB HRR₃ and 41.1 dB HRR₅, in some embodiments. Once improved to 5-bit resolution, the LNTAs now provide 56.7 dB HRR₃ and 61.1 dB HRR₅, in some embodiments. Using a finer resolution or a larger M, in some embodiments, will lead to a higher HRR, in some embodiments. Similarly, in some embodiments, the HRR for M-phase-modulated LNTAs is:

$\begin{matrix} {{HRR}_{i} = {{\frac{{sinc}\left( {\pi/M} \right)}{{sinc}\left( {i\;{\pi/M}} \right)}} \cdot \frac{\sum_{k = 0}^{{M/2} - 1}{{\rho_{{IF},k} \cdot {\cos\left( {\frac{2\pi}{M}k} \right)}}}}{\sum_{k = 0}^{{M/2} - 1}{{\rho_{{IF},k} \cdot {\cos\left( {i\frac{2\pi}{M}k} \right)}}}}}} & (61) \end{matrix}$

where ρ_(IF,k) should be cos(2πk/M), ideally. For the whole signal branch, harmonic rejection happens in both the F_(LO) and the F_(IF) clock domains, in some embodiments. To the first order, its HRR can be obtained by multiplying two HRR expressions, in some embodiments. In some embodiments, for N=M=8, it is:

$\begin{matrix} {{HRR}_{n,m} = {{{\frac{{sinc}\left( {\pi/8} \right)}{{sinc}\left( {m\;{\pi/8}} \right)} \cdot \frac{1 + {2{\rho_{IF} \cdot {\cos\left( {\pi/4} \right)}}}}{1 + {2{\rho_{IF} \cdot {\cos\left( {m\;{\pi/4}} \right)}}}}}} \cdot {{\frac{{sinc}\left( {\pi/8} \right)}{{sinc}\left( {n\;{\pi/8}} \right)} \cdot \frac{1 + {2{\rho_{LO} \cdot {\cos\left( {\pi/4} \right)}}}}{1 + {2{\rho_{LO} \cdot {\cos\left( {n\;{\pi/4}} \right)}}}}}}}} & (62) \end{matrix}$

where ρ_(LO) is the ratio of the baseband weight used in the harmonic recombination for the F_(LO) clock; n and m are the harmonic orders for the F_(LO) and the F_(IF) clocks, respectively. Ideally, both β_(LO) and ρ_(IF) should be cos(π/4), in some embodiments. Note that (62) reduces to (45) when n=k₁N±1, m=k₂M±1, and k1, k2∈

, in some embodiments. This is because the LNTA branches also employ the switching circuits for frequency translations and, thereby, face the same challenges from harmonic folding, in some embodiments.

So far, clocks have been assumed to be ideal with no phase or gain mismatches, and the analog circuits have been assumed to be perfectly phase and gain matched. In practice, non-idealities will occur, and as a result, the low-band outputs will contain signal components that are down-converted from the higher RF carrier and vice versa, in some embodiments.

The model in FIG. 25 is used to study the effects of phase and gain imbalances on the sideband rejection, where γ_(LO) and γ_(IF) are the phase imbalances for the F_(LO) and F_(IF) clock domains, respectively, and ϵ1 and ϵ2 are the amplitude imbalances due to the analog circuitry, in some embodiments. For the quadrature-modulated LNTA branches, the F_(IF) clocks are running at a much lower rate, compared to the F_(LO) clock rate, in some embodiments. Thus, the phase imbalances due to the F_(IF) clocks can be neglected by assuming γ_(IF) is zero, in some embodiments. The sideband rejection is defined as the ratio of the down-converted signal power from the desired RF carrier to the down-converted signal power from the undesired RF carrier:

$\begin{matrix} {{SBR} = {\frac{1 + {\cos\left\lbrack {\gamma_{LO} + {2 \cdot {{atan}\left( {\epsilon_{1}/2} \right)}}} \right\rbrack}}{1 - {\cos\left\lbrack {\gamma_{LO} - {2 \cdot {{atan}\left( {\epsilon_{1}/2} \right)}}} \right\rbrack}}.}} & (63) \end{matrix}$

In some embodiments, to demodulate an uncoded QAM-1024 modulated signal with a bit error ratio of 10⁻⁶, assuming that the received power levels for both carriers are the same, a minimum signal-to-noise ratio (SNR) of 39.0 dB can be used, meaning the phase imbalance can be 1 degree, while the gain imbalances can be below 0.2 dB. Note that the sideband rejection of the double-layer mixer-first branch will share the same expression since its mathematical model is the same as the model in FIG. 25, in some embodiments.

In some embodiments, as M gets large enough (e.g., above 8), the effect of noise folding can be ignored, and the noise performance will be dominated by the noise from the fundamental tone at F_(IF). As M becomes very large, the noise factor in (37) approaches:

$\begin{matrix} {{\lim\limits_{M->\infty}F_{{LB},{N\text{-}M}}^{\prime}} = {\frac{1}{{sinc}^{2}\left( {\pi/N} \right)} \cdot \left( {2 + {\frac{4\gamma}{G_{m,{p\; k}}R_{S}} \cdot \frac{4}{\pi}}} \right)}} & (64) \end{matrix}$

In some embodiments, it is desirable to have a large number (e.g., 16 or above) of LNTA modulation phases for better spurious response profiles, especially when the two carriers are close to each other (i.e., when F_(IF) is small).

In accordance with some embodiments, the double-layer mixer-first branch and the quadrature-modulated LNTA branches are combined to form the proposed double-conversion, noise-cancelling receiver that inherits the input matching properties of the double-layer mixer-first branch while having much better sensitivity.

To derive the Noise-Cancelling (NC) condition, two observations can be made. In some embodiments, the noise of the R_(B) resistors and the baseband op-amps from the mixer-first branch are orthogonal and un-correlated between different signal paths, allowing us to study one path and then generalize its result to all other paths. FIG. 26 can be used to study the NC condition of the (x, y)^(th) path, in some embodiments, where K is the coefficient to adjust the relative gain difference between two branch outputs V_(MFB)(x,y) and V_(LB)(x,y).

Another observation is that, in some embodiments, random noise can be represented as a summation of a great number of equally-spaced sinusoidal tones, whose amplitudes are independent random variables distributed normally about zero, and whose phases are also independent random variables and distributed uniformly from 0 to 2π. Therefore, a noise source can then be replaced with an equivalent AC source, and its AC response within the system can be studied, in some embodiments. Now, its noise source of the R_(B) resistor in the (x, y)^(th) path is replaced with V_(n,RB)(ω), in some embodiments. In some embodiments, its transfer function to V_(MFB)(x,y) is:

$\begin{matrix} {\frac{V_{{MFB}{({x,y})}}(\omega)}{V_{n,R_{B}}(\omega)} = {\frac{R_{F,{MFB}}}{R_{1} + R_{B}} \cdot \frac{1}{1 + {j\;\omega\; R_{F,{MFB}}C_{F,{MFB}}}} \cdot \frac{1 + {j\;\omega\; R_{1}C_{B}}}{1 + {j\;{\omega\left( R_{1}||R_{B} \right)}C_{B}}} \cdot {\cos\left( {{\frac{2\pi}{N}x} - {\frac{2\pi}{M}y}} \right)}}} & (65) \end{matrix}$

where the cos factor stems from harmonic recombination and sideband separation. In some embodiments, the transfer function to V_(BB)(x,y) path is:

$\begin{matrix} {\frac{V_{{BB}{({x,y})}}(\omega)}{V_{n,R_{B}}(\omega)} = {\frac{R_{1}}{R_{1} + R_{B}} \cdot \frac{1}{1 + {j\;{\omega\left( R_{1}||R_{B} \right)}C_{B}}}}} & (66) \end{matrix}$

where R₁=(R_(S)+2R_(SW))·(NM/2). From time-domain, in some embodiments, this voltage will appear at RF input during two time windows, ξ_(x)(t)·ρ_(y)(t) and ξ_(x+N/2)(t)·ρ_(y+M/2)(t). In some embodiments, from frequency-domain, it means that V_(BB)(x,y) will be translated to (pF_(LO)+qF_(IF)) as follows:

$\begin{matrix} {{V_{{RF}{({x,y})}}(\omega)} = {{\mathcal{F}\left\langle {\left\{ {\frac{R_{S}}{R_{S} + {2R_{SW}}} \cdot \left\lbrack {{{\xi_{x}(t)} \cdot {\rho_{y}(t)}} + {{\xi_{x + {N/2}}(t)} \cdot {\rho_{y + {M/2}}(t)}}} \right\rbrack} \right\} \cdot {v_{BB}(t)}} \right\rangle} = {\frac{R_{S}}{R_{S} + {2R_{SW}}} \cdot {\sum\limits_{p = {- \infty}}^{+ \infty}{\sum\limits_{q = {- \infty}}^{+ \infty}{\alpha_{p}{\beta_{q} \cdot \left\{ {1 + {\exp\left\lbrack {{- {j\left( {p + q} \right)}}\pi} \right\rbrack}} \right\} \cdot {\exp\left( {{- {jxp}}\frac{2\pi}{N}} \right)} \cdot {\exp\left( {{- {jyq}}\frac{2\pi}{M}} \right)} \cdot {V_{BB}\left\lbrack {\omega - \left( {{p\;\omega_{LO}} + {q\;\omega_{IF}}} \right)} \right\rbrack}}}}}}}} & (67) \end{matrix}$

where (p+q) is even. This RF voltage is then seen by the modulated LNTA branches, in some embodiments. Since only the baseband components are of interest, in some embodiments, the derivations can be greatly simplified with V_(LB)(x,y) given below:

$\begin{matrix} {{V_{{LB}{({x,y})}}(\omega)} = {{{V_{{BB}{({x,y})}}(\omega)} \cdot \frac{R_{S}}{R_{S} + {2R_{SW}}} \cdot \frac{G_{m,{p\; k}}R_{F,{LB}}}{1 + {j\;\omega\; R_{F,{LB}}C_{F,{LB}}}} \cdot \cos \cdot \left( {{\frac{2\pi}{N}x} - {\frac{2\pi}{M}y}} \right) \cdot {\sum\limits_{p = {- \infty}}^{+ \infty}{\sum\limits_{q = {- \infty}}^{+ \infty}{{\alpha_{p}}^{2}{{\beta_{q}}^{2} \cdot \left\{ {1 + {\exp\left\lbrack {{- {j\left( {p + q} \right)}}\pi} \right\rbrack}} \right\}^{2}}}}}} = {{{V_{n,R_{B}}(\omega)} \cdot G_{m,{p\; k}}}{R_{S} \cdot \frac{R_{F,{LB}}}{R_{1} + R_{B}} \cdot \frac{1}{1 + {j\;{\omega\left( R_{1}||R_{B} \right)}C_{B}}} \cdot \frac{1}{1 + {j\;\omega\; R_{F,{LB}}C_{F,{LB}}}} \cdot {\cos\left( {{\frac{2\pi}{N}x} - {\frac{2\pi}{M}y}} \right)}}}}} & (68) \end{matrix}$

In some embodiments, if attention is further restricted to the frequency components well within the channel (i.e., Δω≈0), V_(CMB(x,y))(ω) becomes:

$\begin{matrix} {\frac{V_{{CMB}{({x,y})}}({\Delta\omega})}{V_{n,R_{B}}({\Delta\omega})} = {\frac{{G_{m,{p\; k}}R_{S}R_{F,{LB}}} + {K \cdot R_{F,{MFB}}}}{R_{1} + R_{B}} \cdot {\cos\left( {{\frac{2\pi}{N}x} - {\frac{2\pi}{M}y}} \right)}}} & (69) \end{matrix}$

where the NC condition can be found by setting (69) to zero as:

$\begin{matrix} {K_{NC}^{\prime} = {{{- 1} \cdot G_{m,{p\; k}}}{R_{S} \cdot \frac{R_{F,{LB}}}{R_{F,{MFB}}}}}} & (70) \end{matrix}$

In some embodiments, following the same logic, the noise source of the baseband op-amp can be replaced with an AC source V_(n,op)(ω) and find its transfer function to V_(CMB(x,y)) as:

$\begin{matrix} {\frac{V_{{CMB},{{OP}{({x,y})}}}({\Delta\omega})}{V_{n,{op}}({\Delta\omega})} = {\frac{1}{R_{1} + R_{B}} \cdot {\cos\left( {{\frac{2\pi}{N}x} - {\frac{2\pi}{M}y}} \right)} \cdot {\quad\left\lbrack {{G_{m,{p\; k}}R_{S}R_{F,{LB}}} + {K \cdot \left( {R_{1} + R_{B} + R_{F,{MFB}}} \right)}} \right\rbrack}}} & (71) \end{matrix}$

While the noise of R_(B) and baseband op-amps is anti-correlated at V_(LB(x,y)) and V_(MFB(x,y)), in some embodiments, the down-converted signals are actually correlated at these two nodes. In some embodiments, under the NC condition, the desired signals add up constructively; the receiver conversion gain becomes:

$\begin{matrix} \begin{matrix} {{CG}_{{RX},{N\text{-}M}}^{\prime} \equiv {{CG}_{{LB},{N\text{-}M}}^{\prime} - {K_{{NC},{RB}}^{\prime} \cdot {CG}_{{MFB},{N\text{-}M}}^{\prime}}}} \\ {= {{CG}_{{LB},{N\text{-}M}}^{\prime} \cdot \left( {1 + \frac{R_{S}}{{2R_{SW}} + {2\eta\; R_{B}}}} \right)}} \end{matrix} & (72) \end{matrix}$

where, as expected, the expression is a function of R_(B). When the input impedance is matched (i.e., 2R_(SW)+2ηR_(B)=R_(S)), the receiver conversion gain is twice the conversion gain of the modulated LNTA branches, in some embodiments.

Regarding the noise from R_(S): Behaving similarly as signals, the noise of R_(S) propagates to the outputs of the two signal branches creating correlated components, in some embodiments. In some embodiments, its noise contribution at V_(CMB) is:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},R_{S}}^{2}}}{\Delta\; f} = {\frac{\overset{\_}{v_{n,R_{S}}^{2}}}{\Delta\; f} \cdot \frac{2}{{{sinc}^{2}\left( {\pi/M} \right)} \cdot {{sinc}^{2}\left( {\pi/N} \right)}} \cdot \left\lbrack {{\frac{{2R_{SW}} + {2\eta\; R_{B}}}{R_{S} + {2R_{SW}} + {2\eta\; R_{B}}} \cdot {CG}_{{LB},{N\text{-}M}}^{\prime}} - {K \cdot {CG}_{{MFB},{N\text{-}M}}^{\prime}}} \right\rbrack_{-}^{2}}} & (73) \end{matrix}$

where the multiplication factor inside the bracket is the voltage division ratio between R_(S) and R_(in). Under the NC condition, (73) reduces to:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},R_{S}}^{2}}}{\Delta f} = {\frac{\overset{\_}{v_{n,R_{S}}^{2}}}{\Delta f} \cdot {CG}_{{LB},{N - M}}^{\prime^{2}} \cdot {\frac{2}{{{sinc}^{2}\left( {\pi/M} \right)} \cdot {{sinc}^{2}\left( {\pi/N} \right)}}.}}} & (74) \end{matrix}$

The noise of R_(S) is now not a function of input matching anymore, in some embodiments. Qualitatively, with large R_(in), the noise at the LNTA branch outputs increases, whereas the noise at the mixer-first branches decreases, and vice versa for the case of small R_(in), in some embodiments. In some embodiments, for both scenarios, the output noise due to R_(S) stays constant.

Regarding the noise from R_(SW): The noise transfer function of R_(SW) is different from that of R_(S), since its noise creates anti-correlated components at two branch outputs, in some embodiments. In some embodiments, its noise contribution at V_(CMB) is given in as follows:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},R_{SW}}^{2}}}{\Delta f} = {2 \cdot \frac{\overset{\_}{v_{n,R_{SW}}^{2}}}{\Delta f} \cdot \left\lbrack {\frac{R_{S} \cdot {CG}_{{LB},{N - M}}^{\prime}}{R_{S} + {2R_{SW}} + {2{\eta R}_{B}}} + {K \cdot \frac{\left( {{2R_{SW}} + {2{\eta R}_{B}}} \right) \cdot {CG}_{{MFB},{N - M}}^{\prime}}{R_{S} + {2R_{SW}} + {2{\eta R}_{B}}}}} \right\rbrack^{2} \cdot \frac{2}{{{sinc}^{2}\left( {\pi/N} \right)} \cdot {{sinc}^{2}\left( {\pi/M} \right)}}}} & (75) \end{matrix}$

In some embodiments, under the NC condition, the noise of R_(SW) gets cancelled, meaning that v_(no,R) _(SW) ² /Δf=0.

Regarding the Noise from R_(B) and baseband op-amps: In some embodiments, the noise transfer functions of the (x, y)^(th) path to V_(CMB)(x,y) are derived above for both R_(B) and the baseband op-amps. In some embodiments, here, the noise contribution at V_(CMB) from R_(B) in all the paths is derived as:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},R_{B}}^{2}}}{\Delta f} = {{\sum\limits_{x = 0}^{{N/2} - 1}{\sum\limits_{y = 0}^{M - 1}{\frac{\overset{\_}{v_{n,R_{B}}^{2}}}{\Delta f} \cdot {\frac{V_{{CMB}{({x,y})}}({\Delta\omega})}{V_{n,R_{B}}({\Delta\omega})}}^{2}}}} = {\frac{\overset{\_}{v_{n,R_{B}}^{2}}}{\Delta f} \cdot \frac{NM}{4} \cdot \left( \frac{{G_{m,{pk}}R_{S}R_{F,{LB}}} + {K \cdot R_{F,{MFB}}}}{R_{1} + R_{B}} \right)^{2}}}} & (76) \end{matrix}$

where its value reduces to zero under the NC condition. In some embodiments, the noise contribution at V_(CMB) from all baseband op-amps in all the paths is:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},{op}}^{2}}}{\Delta f} = {{\sum\limits_{x = 0}^{{N/2} - 1}{\sum\limits_{y = 0}^{M - 1}{\frac{\overset{\_}{v_{n,{op}}^{2}}}{\Delta f} \cdot {\frac{V_{{CMB},{{OP}{({x,y})}}}({\Delta\omega})}{V_{n,{op}}({\Delta\omega})}}^{2}}}} = {\frac{\overset{\_}{v_{n,{op}}^{2}}}{\Delta f} \cdot \frac{NM}{4} \cdot \left( \frac{{G_{m,{pk}}R_{S}R_{F,{LB}}} + {K \cdot \left( {R_{1} + R_{B} + R_{F,{MFB}}} \right)}}{R_{1} + R_{B}} \right)^{2}}}} & (77) \end{matrix}$

where, under the NC condition, its value reduces to:

$\begin{matrix} {\frac{\overset{\_}{v_{{no},{op}}^{2}}}{\Delta f} = {\frac{\overset{\_}{v_{n,{op}}^{2}}}{\Delta f} \cdot \frac{NM}{2} \cdot \left( {{CG}_{{LB},{N - M}}^{\prime} \cdot \frac{R_{S}}{R_{MFB}}} \right)^{2} \cdot \frac{2}{{{sinc}^{2}\left( {\pi/N} \right)} \cdot {{sinc}^{2}\left( {\pi/M} \right)}}}} & (78) \end{matrix}$

where it can now be compared with the noise contribution due to the modulated LNTAs in (55). In some embodiments, using N=M=8, F_(LO)=700 MHz, F_(IF)=150 MHz, R_(S)=50Ω, R_(SW)=10Ω, R_(B)=1.06 kΩ, R_(F,MFB)=15 kΩ, R_(F,LB)=5 kΩ, and Gm,pk=90 mS, (55) can be calculated as 14.2 fV²/Hz, whereas (78) can be calculated as 0.6 fV²/Hz. The noise due to the baseband op-amps under the NC condition is then much smaller than the noise due to the modulated LNTAs and can be ignored, in some embodiments.

Regarding the noise factor of the double-conversion receiver: Since the noise due to the modulated LNTAs stays the same, in some embodiments, the noise factor of the complete receiver, F_(RX,N-M)′, is derived as follows:

$\begin{matrix} {F_{{RX},{N - M}}^{\prime} = {\frac{1}{{{sinc}^{2}\left( {\pi/N} \right)} \cdot {{sinc}^{2}\left( {\pi/M} \right)}} \cdot {\quad\left\lbrack {1 + {{\frac{\gamma}{G_{m,{pl}}R_{S}} \cdot \frac{4}{M}}{\sum\limits_{k = 0}^{{M/2} - 1}{{\cos\left( {\frac{2\pi}{M}k} \right)}}}} + {\frac{\gamma}{G_{m,{op}}R_{S}} \cdot \frac{NM}{2} \cdot \left( \frac{R_{S}}{R_{F,{MFB}}} \right)^{2}}} \right\rbrack}}} & (79) \end{matrix}$

where the third term stems from the noise due to baseband op-amps and is much smaller than the noise due to the modulated LNTAs. In some embodiments, for N=M=8, (79) reduces to:

$\begin{matrix} {F_{{RX},{S - S}}^{\prime} \approx {\frac{1}{{sinc}^{4}\left( {\pi/8} \right)} \cdot \left\{ {1 + {\frac{2\gamma}{G_{m,{pk}}R_{S}} \cdot \left\lbrack {\frac{1}{2} + {\cos\left( \frac{\pi}{4} \right)}} \right\rbrack}} \right\}}} & (80) \end{matrix}$

In some embodiments, similarly, the noise factor of the receiver, F_(RX,8-8)″, in a fully differential realization can be derived as:

$\begin{matrix} {F_{{RX},{S - S}}^{''} \approx {\frac{1}{{sinc}^{4}\left( {\pi/8} \right)} \cdot \left\{ {1 + {\frac{2\gamma}{G_{m,{pk}}R_{S}} \cdot \left\lbrack {\frac{1}{2} + {\cos\left( \frac{\pi}{4} \right)}} \right\rbrack}} \right\}}} & (81) \end{matrix}$

The tuned RF input interface offers attenuation to out-of-band (OB) blocking signals and reduces their voltage swings at the receiver's input (i.e., at the input of the quadrature-modulated LNTA branches) for better blocker tolerance, in some embodiments. In some embodiments, for OB frequencies, R_(SW) dominates the input impedance; the attenuation of a single-ended-differential realization is:

$\begin{matrix} {{ATTN} = {{- 20} \cdot {\log_{10}\left( {2 \cdot \frac{2R_{SW}}{{2R_{SW}} + R_{S}}} \right)}}} & (82) \end{matrix}$

For R_(S)=50Ω and R_(SW)=5Ω, the attenuation is about 10 dB; the voltage swings at the receiver's input are three times smaller than that in a receiver with a broadband termination, in some embodiments. Thus, if the broadband-terminated receiver has a B1 dB of −10 dBm, the receiver with the tuned RF interface is expected to have a B1 dB of 0 dBm, in some embodiments. In some embodiments, for R_(SW)=10Ω, the attenuation reduces to only 5 dB and the B1 dB is improved to −5 dBm. Thus, it is highly desirable to have a small R_(SW), while still maintaining the least amount of parasitic capacitance at the RF input, in some embodiments.

Although the invention has been described and illustrated in the foregoing illustrative embodiments, it is understood that the present disclosure has been made only by way of example, and that numerous changes in the details of implementation of the invention can be made without departing from the spirit and scope of the invention, which is limited only by the claims that follow. Features of the disclosed embodiments can be combined and rearranged in various ways. 

What is claimed is:
 1. A circuit for a receiver, comprising: N first mixers that each receive an input signal, that are each clocked by a different phase of a first common clock frequency, and that each provide an output, wherein N is a count of the first mixers; and for each of the N first mixers: a set of M second mixers, wherein M is a count of the second mixers in the set, wherein each second mixer in the set of M second mixers receives as an input the output of a same one of the N first mixers unique to the set, wherein each of the M second mixers in the set is clocked by a different phase of a second common clock frequency, and wherein each of the second mixers has an output; a set of M resistors having a first side and a second side, wherein the first side of each of the set of M resistors is connected to the output of a corresponding one of the set of M second mixers; and a set of M trans-impedance amplifiers that each having an input connected to the second side of a corresponding one of the set of M resistors and having an output.
 2. The circuit of claim 1, further comprising: a plurality of low noise transconductance amplifier branches each comprising: a transconductor having an input connected to the input signal and a transconductor output signal; N third mixers that each receive a corresponding one of the transconductor output signals, that are each clocked by a different phase of the first common clock frequency, and that each provide a third mixer output signal; and N filters that each receive a corresponding on of the third mixer output signals and provide a filtered output signal.
 3. The circuit of claim 2, wherein the transconductor comprises a plurality of transconductor unit cells that individually controllable.
 4. The circuit of claim 3, wherein each of the plurality of low noise transconductance amplifier branches further comprises a direct digital synthesis circuit the controls the plurality of transconductor unit cells.
 5. The circuit of claim 4, wherein the direct digital synthesis circuit comprises: a numerically controllable oscillator; and an accumulator.
 6. The circuit of claim 1, further comprising a harmonic recombination circuit.
 7. The circuit of claim 6, further comprising a sideband separation circuit.
 8. The circuit of claim 1, wherein each of the first mixers is differential and clocked by two phases at the first common clock frequency each having a 12.5% duty cycle.
 9. The circuit of claim 8, wherein each of the second mixers is differential and clocked by two phases at the second common clock frequency each having a 12.5% duty cycle.
 10. The circuit of claim 1, further comprising, for each of the N first mixers, a set of M capacitors each having a first side and a second side, wherein, for each of the set of M capacitors, the first side of the capacitor is connected to the first side of a corresponding one of the set of M resistors and wherein the second side of the capacitor is connected to a voltage level. 